Semiconductor integrated circuit

ABSTRACT

In a dynamic flip-flop circuit with a data selection function, for example, when data having an H value has been selected using a selection signal S 0 , a first node N 1  is L and a second node N 2  of a second dynamic circuit  1 B is H, so that an output signal Q has an H level. In this case, when none of a plurality of pieces of data D 0  to D 2  is selected using selection signals S 0  to S 2 , the first node N 1  is H, so that the electric charge of the second node N 2  is discharged and the output signal Q erroneously has an L level. However, in this case, an output node N 3  is H and a fourth node N 4  is L, so that an n-type transistor Tr 6  of the second dynamic circuit  1 B is turned OFF, thereby preventing the second node N 2  from being discharged. Therefore, a normal operation is performed while securing a satisfactorily high-speed operation even when none of the pieces of data is selected. This circuit is used in a predetermined circuit, such as, for example, a forwarding path of a data path, a crossbar bus switch, or an input portion of a reconfigurable processing unit.

RELATED APPLICATIONS

This application is a continuation of U.S. patent application Ser. No.12/430,469, filed on Apr. 27, 2009, now U.S. Pat. No. 7,859,310, whichis a divisional of U.S. patent application Ser. No. 11/792,777, filedJun. 11, 2007, now U.S. Pat. No. 7,541,841, which is a U.S. NationalPhase under 35 U.S.C. §371 of International Application No.PCT/JP2006/320637, filed on Oct. 17, 2006, which in turn claims thebenefit of Japanese Application No. 2005-302554, filed on Oct. 18, 2005,the entire contents of each of which are hereby incorporated byreference.

TECHNICAL FIELD

The present invention relates to a semiconductor integrated circuit, andmore particularly, to a high-speed semiconductor integrated circuit.

BACKGROUND ART

Conventionally, the speed of a semiconductor integrated circuit,particularly a flip-flop circuit, is increased by incorporating adynamic circuit into its internal structure as described in, forexample, Patent Document 1. The dynamic flip-flop circuit described inPatent Document 1 has a function of receiving a plurality of pieces ofdata, selecting any one of them, and holding and outputting the selecteddata.

Hereinafter, the structure of the flip-flop circuit having the dataselection function will be described with reference to FIG. 3( a). InFIG. 3( a), a data selection circuit 91 is provided at the previousstage of a holding circuit 90. In the data selection circuit 91, when aclock CLK is at a Low level (Low period), a node N1 is precharged to apower source potential Vdd by a p-type transistor Tr1, while a node N2is precharged to the power source potential Vdd by a p-type transistorTr50. Near the end of this period, one of selection signals S0 to S2which is used to select any one of a plurality of pieces of data D0 toD2 is turned High. Subsequently, when the clock CLK goes to High and theselected data (e.g., D0) is High, the electric charge of the node N1 isdischarged via an n-type transistor Tr2, so that the potential of thenode N1 becomes equal to that of the ground. Therefore, an n-typetransistor Tr51 is turned OFF, so that the precharge potential of thenode N2 is held. In this case, this potential is held as an H value bythe holding circuit 90, which in turn outputs an output signal Q havingthe H value.

On the other hand, when the selected data D0 is Low, the electric chargeof the node N1 is not discharged, so that the potential of the node N1is held as it is the precharge potential and the n-type transistor Tr51is turned ON. As a result, the electric charge of the node N2 isdischarged via the n-type transistor Tr51 and the n-type transistor Tr2,so that the potential of the node N2 becomes an L value. The L value isheld by the holding circuit 90, which in turn outputs an output signal Qhaving the L value.

Note that, in FIG. 3( a), SI indicates a data input when scanning isperformed, SE indicates a scan shift control signal, and SEB indicatesan inverted signal of the scan shift control signal.

Patent Document 1: Japanese Unexamined Patent Application PublicationNo. 2003-060497

DISCLOSURE OF THE INVENTION Problems to be Solved by the Invention

However, it was found that the conventional dynamic flip-flop circuithaving the data selection function malfunctions when none of theplurality of pieces of data is selected. Hereinafter, the malfunctionwill be described.

In an ordinary operation, for example, the node N2 is at the prechargepotential (H value) and the holding circuit 90 outputs the output signalQ having the H value. In this case, when none of the plurality of piecesof data D0 to D2 is selected during the next High period of the clockCLK (i.e., all of the selection signals S0 to S2 have the Low value),the n-type transistor Tr2 is turned ON. However, the precharge potentialof the node N1 is held, so that the n-type transistor Tr51 is turned ON.Therefore, the electric charge of the node N2 is discharged via then-type transistors Tr51 and Tr2 to the L value. As a result, the holdingcircuit 90 erroneously outputs an output signal Q having the L value.

To solve the above-described problems, for example, the followingcircuit is considered to be added, which inputs a signal to the gate ofthe n-type transistor Tr2 as shown in FIG. 3( b). Specifically, a staticcircuit comprising a circuit 92 including an OR circuit which receivesall of the selection signals S0 to S2 and a latch circuit which latchesan output of the OR circuit during a High period of the clock CLK, andan AND circuit 93 which receives an output of the latch circuit and theclock CLK, is additionally provided, and an output of the AND circuit 93is input to the gate of the n-type transistor Tr2.

In this case, however, all of the selection signals S0 to S2 need to bepassed through the OR circuit and the latch circuit by a rising time ofthe clock CLK. Therefore, an extra setup time (a time required for thestatic circuit to establish its output by a rising time of the clockCLK) is required, resulting in impairment of the speed of operation.

In view of the problems above, an object of the present invention is toprovide a dynamic flip-flop circuit with a data selection function whichcan operate normally while securing a satisfactorily high-speedoperation even when none of a plurality of pieces of data is selected.

Also, the area of the dynamic flip-flop circuit with a data selectionfunction is decreased by reducing the number of circuit elements.Further, by providing a high-speed and small-area dynamic flip-flopcircuit with a data selection function at an optimal portion, theperformance of a semiconductor integrated circuit is caused to havehigher precision.

Solution to the Problems

To achieve the object, according to the present invention, when none ofa plurality of pieces of data is selected, for example, in thesemiconductor integrated circuit of FIG. 3( a), the precharge of thenode N2 is prevented from being discharged, so that the H value of thenode N2 is maintained. The holding circuit holds and outputs the H valueof the node N2.

Specifically, according to the present invention, in a semiconductorintegrated circuit which receives a clock, a plurality of pieces ofdata, and a plurality of selection signals for selecting the data, andwhen the clock is transitioned, outputs a piece of data selected by theselection signal to a holding circuit, the semiconductor integratedcircuit comprises a non-selected state detection circuit for detecting astate in which all of the plurality of selection signals select none ofthe plurality of pieces of data. When the non-selected state detectioncircuit detects a state in which all of the plurality of selectionsignals select none of the plurality of pieces of data, the previouslyselected data is prevented from being changed, so that output data ofthe holding circuit is held. The semiconductor integrated circuit isused in a predetermined circuit.

In the semiconductor integrated circuit of the present invention, thepredetermined circuit is a forwarding path of a data path.

In the semiconductor integrated circuit of the present invention, thepredetermined circuit is a crossbar bus switch.

In the semiconductor integrated circuit of the present invention, thepredetermined circuit is an input portion of a reconfigurable processingunit.

The semiconductor integrated circuit of the present invention has afirst dynamic circuit for receiving the plurality of pieces of data andthe plurality of selection signals, and selecting any of the pluralityof pieces of data based on the plurality of selection signals, a seconddynamic circuit for receiving an output of the first dynamic circuit,and a differential amplification circuit for receiving the output of thefirst dynamic circuit and an output of the second dynamic circuit, andamplifying a differential voltage between the inputs, the differentialamplification circuit being activated by the clock. An output of thedifferential amplification circuit is input to the holding circuit.

The semiconductor integrated circuit of the present invention has afirst dynamic circuit for receiving the plurality of pieces of data andthe plurality of selection signals, and selecting any of the pluralityof pieces of data based on the plurality of selection signals, a seconddynamic circuit for receiving an output of the first dynamic circuit, athird dynamic circuit for receiving the plurality of selection signals,and determining whether or not any of the plurality of selection signalsis activated, a fourth dynamic circuit for receiving an output of thethird dynamic circuit, and a differential amplification circuit foramplifying a differential voltage between the output of the firstdynamic circuit and an output of the second dynamic circuit, thedifferential amplification circuit being activated by an output of thefourth dynamic circuit or an inverted output of the third dynamiccircuit. An output of the differential amplification circuit is input tothe holding circuit.

The semiconductor integrated circuit of the present invention has afirst dynamic circuit for receiving the plurality of pieces of data andthe plurality of selection signals, and selecting any of the pluralityof pieces of data based on the plurality of selection signals, thenumber of stages of transistors connected in series so as to select anyof the plurality of pieces of data being a predetermined number. In thenon-selected state detection circuit, the number of stages oftransistors connected in series so as to detect a state in which all ofthe plurality of selection signals select none of the plurality ofpieces of data is smaller, by one or more, than the predetermined numberof stages in the first dynamic circuit.

The semiconductor integrated circuit of the present invention comprisesa first dynamic circuit for receiving the plurality of pieces of dataand the plurality of selection signals, and selecting any of theplurality of pieces of data based on the plurality of selection signals,a second dynamic circuit for receiving an output of the first dynamiccircuit, and a setup absorption circuit for absorbing a setup delay ofdata selected by the plurality of selection signals of the plurality ofpieces of data, the setup absorption circuit being provided between thesecond dynamic circuit and the holding circuit.

In the semiconductor integrated circuit of the present invention, anoutput of the second dynamic circuit is input to the holding circuit.The setup absorption circuit is activated by a delayed clock signalwhich is obtained by delaying the clock by a predetermined time, theoutput of the first dynamic circuit is input to the setup absorptioncircuit, and an output side of the setup absorption circuit is connectedto an output side of the second dynamic circuit via a switch circuitwhich is controlled by the delayed clock signal.

In the semiconductor integrated circuit of the present invention, anoutput of the second dynamic circuit is input to the holding circuit.The setup absorption circuit is activated by an output signal of thesecond dynamic circuit, the output of the first dynamic circuit is inputto the setup absorption circuit, and an output side of the setupabsorption circuit is connected to an output side of the second dynamiccircuit via a buffer and a switch circuit which is controlled by theoutput signal of the second dynamic circuit.

According to the present invention, in a semiconductor integratedcircuit which receives a clock and data, and when the clock istransitioned, outputs the data to a holding circuit, the semiconductorintegrated circuit has a first dynamic circuit for receiving the data, asecond dynamic circuit for receiving an output of the first dynamiccircuit, and a differential amplification circuit for receiving theoutput of the first dynamic circuit and an output of the second dynamiccircuit, and amplifying a differential voltage between the inputs. Anoutput of the differential amplification circuit is input to the holdingcircuit.

According to the present invention, in a semiconductor integratedcircuit which receives a clock, a plurality of pieces of data, and aplurality of selection signals for selecting any of the plurality ofpieces of data, and when the clock is transitioned, outputs a piece ofdata selected by the selection signal to a holding circuit, thesemiconductor integrated circuit has a first dynamic circuit forreceiving the plurality of pieces of data and the plurality of selectionsignals, and selecting any of the plurality of pieces of data based onthe plurality of selection signals, a second dynamic circuit forreceiving an output of the first dynamic circuit, a third dynamiccircuit for receiving the plurality of selection signals, anddetermining whether or not any of the plurality of selection signals isactivated, a fourth dynamic circuit for receiving an output of the thirddynamic circuit, and a differential amplification circuit for amplifyinga differential voltage between the output of the first dynamic circuitand an output of the second dynamic circuit, the differentialamplification circuit being activated by an output of the fourth dynamiccircuit or an inverted output of the third dynamic circuit. An output ofthe differential amplification circuit is input to the holding circuit.

According to the present invention, in a semiconductor integratedcircuit which receives a clock, a plurality of pieces of data, and aplurality of selection signals for selecting any of the plurality ofpieces of data, and when the clock is transitioned, outputs selecteddata to a holding circuit, the semiconductor integrated circuit has afirst dynamic circuit for receiving the plurality of pieces of data andthe plurality of selection signals, and selecting any of the pluralityof pieces of data, a second dynamic circuit for receiving an output ofthe first dynamic circuit, a third dynamic circuit for receiving theplurality of selection signals, and a fourth dynamic circuit forreceiving an output of the third dynamic circuit. Only when an output ofthe fourth dynamic circuit is transitioned, an output of the seconddynamic circuit is input to the holding circuit.

According to the present invention, in a semiconductor integratedcircuit which receives a clock and data, and when the clock istransitioned, outputs the data to a holding circuit, the semiconductorintegrated circuit has a first dynamic circuit for receiving the data, asecond dynamic circuit for receiving an output of the first dynamiccircuit, and a third dynamic circuit activated by a delayed clock signalwhich is obtained by delaying the clock by a predetermined time. Anoutput of the second dynamic circuit is input to the holding circuit. Aninput of the third dynamic circuit is the output of the first dynamiccircuit. An output of the third dynamic circuit is connected to theoutput of the second dynamic circuit via a switch circuit which iscontrolled by the delayed clock signal.

According to the present invention, in a semiconductor integratedcircuit which receives a clock and data, and when the clock istransitioned, outputs the data to a holding circuit, the semiconductorintegrated circuit has a first dynamic circuit for receiving the data, asecond dynamic circuit for receiving an output of the first dynamiccircuit, and a third dynamic circuit activated by an output signal ofthe second dynamic circuit. An output of the second dynamic circuit isinput to the holding circuit. An input of the third dynamic circuit isthe output of the first dynamic circuit. An output of the third dynamiccircuit is connected to the output of the second dynamic circuit via abuffer and a switch circuit which is controlled by the output signal ofthe second dynamic circuit.

As described above, according to the present invention, in a dynamicflip-flop circuit with a data selection function, assuming that anoutput signal of a data selection circuit is, for example, High, evenwhen a state subsequently occurs in which no selection signal isactivated, so that no data is selected, this state is detected, so thatthe output signal of the data selection circuit is held High. Therefore,a flip-flop which does not malfunction, has a small area, and can switchwith high speed, can be provided in a portion where the characteristicshave the greatest effect, resulting in higher precision of performanceof a semiconductor integrated circuit.

Further, assuming that selected data is transferred to and held in theholding circuit, when a differential voltage between the output voltagesof the two dynamic circuits is small, the differential voltage isamplified by the differential amplification circuit with high speed, andthe result is output to the holding circuit. Therefore, it is possibleto increase the speed of transition of the output signal of the holdingcircuit.

In addition, in the non-selected state detection circuit, the number ofstages of transistors connected in series so as to detect a state inwhich none of a plurality of selection signals is selected is smaller,by one or more, than the number of stages of transistors connected inseries so as to select any of a plurality of pieces of data. Therefore,when a semiconductor integrated circuit is physically designed (layoutdesign), a vertically symmetrical layout design can be achieved even ifa cell height is low.

Also, even when there is some setup delay in data of a plurality ofpieces of data selected by a plurality of selection signals, the setupdelay is absorbed by the setup absorption circuit, so that the outputsignal of the holding circuit takes a normal value.

EFFECT OF THE INVENTION

As described above, according to the semiconductor integrated circuit ofthe present invention, in a dynamic flip-flop circuit with a dataselection function, even when a state occurs in which no selectionsignal is activated, so that no data is selected, the output signal ofthe data selection circuit is held at the previous value, so that theoutput signal of the holding circuit can be satisfactorily held at theprevious value.

Further, a flip-flop which has a small area and can switch with highspeed can be provided in an optimal portion, resulting in higherprecision of performance of the semiconductor integrated circuit.

Further, since the differential amplification circuit is provided, it ispossible to increase the speed of transition of the output signal of theholding circuit, resulting in a high-speed operation.

In addition, when the semiconductor integrated circuit is physicallydesigned (layout design), a vertically symmetrical layout design can beachieved even if a cell height is low.

Also, even when there is some setup delay in selected data, the setupdelay is absorbed by the setup absorption circuit, so that the outputsignal of the holding circuit takes a normal value. Therefore, it ispossible to achieve a circuit robust to variations in manufacturingprocess or fluctuations in power source voltage

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a diagram illustrating a semiconductor integrated circuitaccording to Embodiment 1 of the present invention.

FIG. 2 is a diagram illustrating an outline of a layout structure of amajor portion of the semiconductor integrated circuit.

FIG. 3( a) is a diagram illustrating a major structure of a conventionalsemiconductor integrated circuit. FIG. 3( b) is a diagram illustrating aproposed example which removes a drawback of the semiconductorintegrated circuit.

FIG. 4 is a diagram illustrating an operation timing chart of thesemiconductor integrated circuit of Embodiment 1 of the presentinvention.

FIG. 5 is a diagram illustrating an internal structure of an outputcircuit included in a semiconductor integrated circuit according toEmbodiment 2 of the present invention.

FIG. 6 is a diagram illustrating an internal structure of a circuit ofgenerating a clock to be supplied to the output circuit.

FIG. 7 is a diagram illustrating an operation timing chart of the outputcircuit and the clock generation circuit.

FIG. 8 is a diagram illustrating a structure of a semiconductorintegrated circuit according to Embodiment 3 of the present invention.

FIG. 9 is a diagram illustrating a variation of the semiconductorintegrated circuit of FIG. 1.

FIG. 10 is a diagram illustrating a layout structure of a major portionof the semiconductor integrated circuit of FIG. 9.

FIG. 11 is a diagram illustrating another variation of the semiconductorintegrated circuit of FIG. 9.

FIG. 12 is a diagram illustrating a structure of a semiconductorintegrated circuit according to Embodiment 4 of the present invention.

FIG. 13 is a timing chart of each node with respect to a signal inputpattern in the semiconductor integrated circuit of Embodiment 4.

FIG. 14 is a timing chart of each node with respect to another signalinput pattern in the semiconductor integrated circuit of Embodiment 4.

FIG. 15 is a diagram illustrating a structure of a semiconductorintegrated circuit according to Embodiment 5 of the present invention.

FIG. 16 is a timing chart of each node with respect to a signal inputpattern in the semiconductor integrated circuit of Embodiment 5.

FIG. 17 is a timing chart of each node with respect to another signalinput pattern in the semiconductor integrated circuit of Embodiment 5.

FIG. 18 is a timing chart of each node with respect to still anothersignal input pattern in the semiconductor integrated circuit ofEmbodiment 5.

FIG. 19 is a diagram illustrating a structure of a semiconductorintegrated circuit according to Embodiment 6 of the present invention.

FIG. 20 is a diagram illustrating a structure of a semiconductorintegrated circuit according to Embodiment 7 of the present invention.

FIG. 21 is a diagram illustrating a structure of a variation of asemiconductor integrated circuit according to Embodiment 7 of thepresent invention.

FIG. 22 is a diagram illustrating a structure of a semiconductorintegrated circuit according to Embodiment 8 of the present invention.

FIG. 23 is a diagram illustrating another static flip-flop according toEmbodiment 8.

FIG. 24 is a diagram illustrating still another static flip-flopaccording to Embodiment 8.

FIG. 25 is a diagram illustrating a data path according to Embodiment 9.

FIG. 26 is a diagram illustrating a crossbar bus switch according toEmbodiment 10.

FIG. 27 is a diagram illustrating a reconfigurable processor accordingto Embodiment 11.

FIG. 28 is a diagram illustrating a structure of a semiconductorintegrated circuit according to Embodiment 12.

FIG. 29 is a diagram illustrating another structure of the semiconductorintegrated circuit of Embodiment 12.

FIG. 30 is a diagram illustrating a semiconductor integrated circuitaccording to Embodiment 13.

FIG. 31 is a diagram illustrating a structure of a semiconductorintegrated circuit according to Embodiment 14.

FIG. 32 is a diagram illustrating a structure of a semiconductorintegrated circuit according to Embodiment 15.

FIG. 33 is a diagram illustrating a timing chart of these miconductorintegrated circuit of Embodiment 15.

FIG. 34 is a diagram illustrating another structure of a setupabsorption circuit included in the semiconductor integrated circuit ofEmbodiment 15.

DESCRIPTION OF THE REFERENCE CHARACTERS

1A first dynamic circuit of NOR type

2A first dynamic circuit of NOR type

1B second dynamic circuit of NAND type

1C third dynamic circuit of NOR type (non-selected state detectioncircuit)

2C third dynamic circuit of NOR type (matching detection circuit)

1D fourth dynamic circuit of NAND type

1E output circuit

Tr20 first p-type transistor

Tr21 first n-type transistor

Tr22 second n-type transistor

1F holding circuit

IN5 first inverter circuit

IN6 second inverter circuit

Tr27 first p-type transistor

Tr28 first n-type transistor

Tr29 second n-type transistor

1G output circuit

70 differential circuit

71 OR circuit

Tr36 n-type transistor (control transistor)

Tr37 n-type transistor (resistance element)

1H clock generation circuit (signal generation circuit)

75 short pulse generation circuit

NAND1 NAND circuit

BEST MODE FOR CARRYING OUT THE INVENTION

Hereinafter, embodiments of the present invention will be described withreference to the accompanying drawings.

Embodiment 1

FIG. 1 illustrates a semiconductor integrated circuit according toEmbodiment 1 of the present invention.

In FIG. 1, D0, D1 and D2 indicate data; S0, S1 and S2 indicate selectionsignals which are used to select the data D0 to D2, respectively; SIindicates a data input when scanning is performed; SE indicates a scanshift control signal which is used to perform a scan shift operation;SEB indicates an inverted signal of the scan shift control signal; and Qand SO indicate output terminals.

The semiconductor integrated circuit of FIG. 1 has a first dynamiccircuit 1A of NOR type, a second dynamic circuit 1B of NAND type, athird dynamic circuit 1C of NOR type, a fourth dynamic circuit 1D ofNAND type, an output circuit 1E, and a holding circuit 1F. The outputcircuit 1E and the holding circuit 1F constitute a dynamic flip-flopcircuit.

The first dynamic circuit 1A of NOR type receives the three pieces ofdata D0 to D2, the three selection signals S0 to S2, and a first clockCLK1, and has two p-type MOS transistors Tr1 and Tr3 and an n-type MOStransistor Tr2.

The first dynamic circuit 1A controls the selection signals S0 to S2 tobe all Low during a first period which is a half period of the firstclock CLK1 from falling to rising (i.e., the first clock CLK1 is Low).Therefore, during the first period, the p-type transistor Tr1 is ON anda first output node N1 is precharged to a power source voltage Vdd.Thereafter, during a second period which is a half period of the firstclock CLK1 from rising to falling (i.e., the first clock CLK1 is High),the p-type transistors Tr1 and Tr3 are OFF, while the n-type transistorTr2 is ON, so that any one of the selection signals S0 to S2 iscontrolled to be High. Therefore, during the second period, thepotential of the first output node N1 is determined, depending on thevalue of one of the data D0 to D2 which is selected using the selectionsignal which is controlled to be High. For example, when the data D0 isLow, the precharged state of the first output node N1 is held and thefirst output node N1 is held at the power source potential Vdd. On theother hand, when the data D0 is High, the electric charge of the firstoutput node N1 is discharged via the n-type transistor Tr2, so that thepotential of the first output node N1 becomes equal to the groundpotential.

The second dynamic circuit 1B of NAND type receives a second clock CLK2and a signal from the first output node N1 of the first dynamic circuit1A. Further, the second dynamic circuit 1B of NAND type has two p-typeMOS transistors Tr4 and Tr8, and three n-type MOS transistors Tr5 toTr7. The gate of the n-type transistor Tr5 receives a signal of thefirst output node N1 of the first dynamic circuit 1A.

In the second dynamic circuit 1B, during a first period in which thesecond clock CLK2 is Low, the p-type transistor Tr4 is ON and the n-typetransistor Tr7 is OFF. Therefore, in this case, a second output node N2is precharged to the power source potential Vdd. Thereafter, during asecond period in which the second clock CLK2 is High, the prechargeoperation is stopped and the n-type transistor Tr5 is turned ON/OFF,depending on the potential of the first output node N1 of the firstdynamic circuit 1A. For example, when Low data D0 is selected, then-type transistor Tr5 is turned OFF and the precharged state of thesecond output node N2 is held. On the other hand, when High data D0 isselected, the n-type transistor Tr5 is turned ON and the electric chargeof the second output node N2 is discharged via the two n-typetransistors Tr5 and Tr7. The n-type transistor Tr6 is an importanttransistor for Embodiment 1, and a function thereof will be describedbelow.

The third dynamic circuit 1C of NOR type receives a third clock CLK3,the three selection signals S0 to S2, and the scan shift control signalSE, and has two p-type transistors Tr9 and Tr11, an n-type transistorTr10, and an inverter IN3.

In the third dynamic circuit (non-selected state detection circuit) 1C,during a first period in which the third clock CLK3 is Low, the p-typetransistor Tr9 is turned ON and the n-type transistor Tr10 is turnedOFF, so that a third output node N3 is precharged to the power sourcepotential Vdd. Thereafter, during a second period in which the thirdclock CLK3 is High, when all of the three selection signals S0 to S2 andthe scan shift control signal SE are Low (i.e., none of the data D0 toD2 is selected), the precharged state of the third output node N3 isheld and this state is detected. On the other hand, when any one of theselection signals goes to High, the electric charge of the third outputnode N3 is discharged via the n-type transistor Tr10, so that thepotential thereof becomes Low.

Further, the fourth dynamic circuit 1D of NAND type receives a fourthclock CLK4 and a signal of the third output node N3 of the third dynamiccircuit 1C, and has two p-type MOS transistors Tr12 and Tr15 and twon-type MOS transistors Tr13 and Tr14. The gate of the n-type MOStransistor Tr13 receives the signal of the third output node N3 of thethird dynamic circuit 1C.

In the fourth dynamic circuit 1D of NAND type, during a first period inwhich the fourth clock CLK4 is Low, the p-type transistor Tr12 is ON andthe n-type MOS transistor Tr14 is OFF, so that a fourth output node N4is precharged to the power source potential Vdd. On the other hand,during a second period in which the fourth clock CLK4 is High, thep-type transistor Tr12 is OFF, so that the precharge operation isstopped and the n-type MOS transistor Tr14 is ON. Therefore, thepotential of the fourth output node N4 is determined, depending on theON/OFF of the n-type transistor Tr13. In other words, during the secondperiod, the electric charge of the third output node N3 of the thirddynamic circuit 1C is held. In other words, in an ordinary operation,when all of the selection signals S0 to S2 are Low and none of the dataD0 to D2 is selected, the electric charge of the fourth output node N4is discharged via the n-type transistors Tr13 and Tr14, so that thepotential of the fourth output node N4 becomes Low. On the other hand,when High data is selected from any of the selection signals S0 to S2and the electric charge of the third output node N3 of the third dynamiccircuit 1C is discharged, the n-type MOS transistor Tr13 is turned OFF,so that the precharged state of the fourth output node N4 is held.

The second dynamic circuit 1B of NAND type is provided with the n-typeMOS transistor Tr6 which receives via its gate a signal of the fourthoutput node N4 of the fourth dynamic circuit 1D. The n-type transistorTr6 is connected in series to the n-type transistor Tr5. When the n-typetransistor Tr5 is ON and the n-type transistor Tr6 is OFF, the electriccharge of the second output node N2 is not discharged, so that theprecharged state thereof is held.

In this embodiment, in the second dynamic circuit 1B of NAND type, whennone of the data D0 to D2 is selected and the n-type transistor Tr5 isON, the n-type transistor Tr6 needs to be already OFF. To achieve this,the third and fourth dynamic circuits 1C and 1D which control the n-typetransistor Tr6 have a structure which allows a higher-speed operationthan that of the first dynamic circuit 1A. For example, the thirddynamic circuit 1C has two stages of transistors connected in series ona pathway from the third output node N3 to the ground, while the firstdynamic circuit 1A has three stages of transistors connected in serieson a pathway from the first output node N1 to the ground. Therefore, thethird dynamic circuit 1C has a higher operating speed than that of thefirst dynamic circuit 1A. In addition, the third and fourth dynamiccircuits 1C and 1D are disposed nearer the second dynamic circuit 1Bthan the first dynamic circuit 1A. As a result, a delay time requiredfor a change in the potentials of the third and fourth output nodes N3and N4 of the third and fourth dynamic circuits 1C and 1D to bepropagated to the n-type transistor Tr6 of the second dynamic circuit 1Bis reduced to be shorter than a delay time required for a potentialchange of the first output node N1 of the first dynamic circuit 1A to bepropagated to the n-type transistor Tr5 of the second dynamic circuit1B.

Further, in order to cause the third and fourth dynamic circuits 1C and1D to operate with higher speed than that of the first dynamic circuit1A, a voltage supplied to the third and fourth dynamic circuits 1C and1D may be set to be higher than that of the first dynamic circuit 1A;the threshold voltage of an MOS transistor included in the third andfourth dynamic circuits 1C and 1D may be set to be lower than thethreshold voltage of an MOS transistor included in the first dynamiccircuit 1A; or a size of the MOS transistor included in the third andfourth dynamic circuits 1C and 1D may be set to be larger than a size ofthe MOS transistor included in the first dynamic circuit 1A. Further,when an STI (Shallow Trench Isolation region) is formed on thesemiconductor substrate, it is expected that the performance of thetransistor is deteriorated due to an influence of the STI, andtherefore, the arrangement or configuration may be adapted inconsideration of the influence of the STI. For example, as shown in FIG.2, when a transistor series 61 is formed on an N-type substrate 60, aplurality of transistors of the transistor series 61 which arepositioned at an edge thereof, are used to constitute an n-typetransistor of the first dynamic circuit 1A, while a plurality oftransistors of the transistor series 61 which are positioned in aninside thereof, are used to constitute an n-type transistor in the thirdand fourth dynamic circuits 1C and 1D. With this structure, an isolationregion (STI) 65 is provided between the transistor series 61 and othertransistor series 62 and 63 on the N-type substrate 60. Therefore, atransistor located at the edge of the transistor series 61 issignificantly deteriorated due to the influence of the STI. However,this transistor is the n-type transistor of the first dynamic circuit 1Afor which a high operating speed is not required, and therefore, thedeterioration has a less influence. On the other hand, the n-typetransistors of the third and fourth dynamic circuits 1C and 10 for whicha high operating speed is required, are comprised of transistors whichare located inside the transistor series 61 so that they are notsignificantly influenced by the STI. Therefore, the high operating speedcan be satisfactorily secured.

Although, in this embodiment, the third and fourth dynamic circuits 1Cand 10 are constructed to have a higher operating speed than that of thefirst dynamic circuit 1A, the present invention is not limited to this,i.e., this structure is not necessarily adopted. For example, althoughthe second clock CLK2 is input to the gate of the n-type transistor Tr7of the second dynamic circuit 1B in the semiconductor integrated circuitof FIG. 1, an inverted signal of the third output node N3 of the thirddynamic circuit 1C may be input instead of the second clock CLK2. In thecase of this structure, when none of the data is selected (i.e., all ofthe selection signals S0 to S2 are Low) before rising of the fourthclock CLK4, the third output node N3 becomes High, so that the n-typetransistor Tr7 is turned OFF. Thereafter, when the fourth clock CLK4rises, the fourth output node N4 becomes Low, so that the n-typetransistor Tr6 is turned OFF. Therefore, the third and fourth dynamiccircuits 1C and 1D do not have to be constructed so that their operatingspeed is higher than that of the first dynamic circuit 1A.

Next, the output circuit 1E and the holding circuit 1F which are theremaining portion of the dynamic flip-flop circuit will be described.The output circuit 1E receives a signal of the first output node N1 ofthe first dynamic circuit 1A and a signal of the second output node N2of the second dynamic circuit 1B, and comprises an inverter IN4, a NORcircuit NOR1, a first p-type MOS transistor Tr20, and three n-type MOStransistor Tr21, Tr22 and Tr23. The drain of the p-type MOS transistorTr20 and the drain of the first the n-type transistor Tr21 are connectedto each other to form a seventh output node N7. A signal of the secondoutput node N2 of the second dynamic circuit 1B is input to the gate ofthe p-type MOS transistor Tr20. The NOR circuit NOR1 comprises twop-type transistors Tr24 and Tr25 and an n-type transistor Tr26, andreceives a signal of the first output node N1 of the first dynamiccircuit 1A and a signal obtained by inverting a signal of the secondoutput node N2 of the second dynamic circuit 1B using the inverter IN4,and outputs a signal as a sixth output node N6 to the gate of first then-type transistor Tr21.

Therefore, in the output circuit 1E, when the second output node N2 ofthe second dynamic circuit 1B is Low and the first output node N1 of thefirst dynamic circuit 1A is High, the p-type transistor Tr20 is turnedON and the n-type transistor Tr21 is turned OFF, so that the seventhoutput node N7 is precharged to the power source potential Vdd, i.e.,the potential thereof becomes High. On the other hand, when the secondoutput node N2 is High and the first output node N1 is Low, the p-typetransistor Tr20 is turned OFF and the n-type transistor Tr21 is turnedON, so that the electric charge of the seventh output node N7 isdischarged, i.e., the potential thereof becomes Low.

In the output circuit 1E, the gate of the second n-type transistor Tr22receives a signal of the fourth output node N4 of the fourth dynamiccircuit 1D, the source of the second n-type transistor Tr22 is grounded,and the drain of the second n-type transistor Tr22 is connected to thesource of the n-type transistor Tr21. In the n-type transistor Tr22,when the potential of the seventh output node N7 is High, the output ofthe NOR circuit NOR1 (sixth output node N6) becomes High due to areduction in the potential of the first output node N1 of the firstdynamic circuit 1A. In this case, even if the n-type transistor Tr21 isturned ON, since the n-type transistor Tr22 is held in the OFF state,the potential of the seventh output node N7 is prevented fromerroneously becoming Low and a through current is prevented.

Next, the holding circuit 1F will be described. The holding circuit 1Ffunctions as a feedback buffer, and comprises a first inverter IN5 and asecond inverter IN6. The seventh output node N7 of the holding circuit1E is connected to the input side of the first inverter IN5. Theinverter IN5 is connected to the input side of the second inverter IN6.The output side of the second inverter IN6 is connected to the seventhoutput node N7. Further, the holding circuit 1F comprises a secondn-type MOS transistor Tr29 connected in series between a first p-typeMOS transistor Tr27 and a first n-type MOS transistor Tr28 whichconstitute the second inverter IN6, and a delay cell 59. The invertersIN5 and IN6 each hold the potential of the seventh output node N7 of theholding circuit IE. The held potential is inverted by the inverter IN7before being output through the output terminal Q. An output of thefirst inverter IN5 is delayed by a predetermined time in the delay cell59 before being output through the output terminal SO.

In the holding circuit 1F, the gate of the n-type MOS transistor Tr29receives a signal of the second output node N2 of the second dynamiccircuit 1B, the drain of the n-type MOS transistor Tr29 is connected tothe drain of the p-type transistor Tr27, and the source of the n-typeMOS transistor Tr29 is connected to the drain of the n-type transistorTr28. The n-type transistor Tr29 has the following function.Specifically, when the seventh output node N7 of the output circuit 1Eis Low, the output node N7 is grounded via the n-type transistor Tr28 ofthe second inverter IN6. When the second output node N2 of the seconddynamic circuit 1B starts going from High to Low, the p-type transistorTr20 of the output circuit 1E is turned ON, so that the seventh outputnode N7 starts being precharged to the power source potential Vdd. Inthis case, the n-type transistor Tr29 is turned OFF by causing thesecond output node N2 to be Low, so that a pathway from the seventhoutput node N7 via the n-type transistor Tr28 to the ground is cut off,thereby promoting the precharge operation of the seventh output node N7.

Next, an operation of the semiconductor integrated circuit of Embodiment1 will be described with reference to a timing chart illustrated in FIG.4. For the sake of simplicity, it is assumed that first to fourth clocksCLK1 to CLK4 are the same clock CLK.

During a first period of the clock CLK, the data D0 is High in a datavalid period (a time satisfying setup and hold times) before and afterrising of the clock, and after the data valid period has passed, thedata D0 becomes Low. The other data D1 and D2 are High. The selectionsignal S0 is Low during the data valid period and becomes High after thedata valid period has passed. The other selection signals S1 and S2 areLow. Therefore, during the first period, none of the data D0 to D2 isselected.

In this state, during the data valid period, the first output node N1 ofthe first dynamic circuit 1A is High, and therefore, the n-typetransistor Tr5 is turned ON in the second dynamic circuit 1B. As aresult, in the conventional example of FIG. 3( a), when the secondoutput node N2 is High, the second output node N2 erroneously goes toLow, so that the flip-flop circuit erroneously outputs an “L” signalinstead of a true “H” signal.

However, in this embodiment, the third output node N3 of the thirddynamic circuit 1C is High, and the fourth output node N4 of the fourthdynamic circuit 1D becomes Low after rising of the clock. Therefore, inthe second dynamic circuit 1B, the n-type transistor Tr6 is turned OFFbefore the n-type transistor Tr5 is turned ON, so that the second outputnode N2 is prevented from erroneously becoming Low, i.e., the secondoutput node N2 is held High. Therefore, in the output circuit 1E, theseventh output node N7 is held Low, so that the holding circuit 1Foutputs the true “H” signal.

On the other hand, it is now assumed that the seventh output node N7 ofthe output circuit 1E is held High. For example, even if the selectionsignal S2 becomes High after rising of the clock CLK and the firstoutput node N1 of the first dynamic circuit 1A becomes Low (not shown),the sixth output node N6 becomes High in the output circuit 1E, so thatthe n-type transistor Tr21 is turned ON. In this case, however, then-type transistor Tr22 is turned OFF, so that the seventh output node N7is not grounded, so that the seventh output node N7 is held High. Thisis because, in the OFF operation of the n-type transistor Tr22, evenwhen the third output node N3 of the third dynamic circuit 1C becomesLow as the selection signal S2 goes to High, the fourth output node N4of the fourth dynamic circuit 1D is held Low.

Note that FIG. 4 illustrates that the data D0 is Low, the data D1 and D2are High, the selection signal S0 is High, and the other selectionsignals S1 and S2 are Low, i.e., the data D0 is selected, during thesecond period of the clock CLK.

In this embodiment, an OR circuit or a latch circuit is not providedbefore the clock as illustrated in FIG. 3( b), and therefore, it is notnecessary to set up a selection signal, thereby making it possible toprovide a dynamic flip-flop circuit capable of operating with highspeed.

Although, in the above description about the operation, the first tofourth clocks CLK1 to CLK4 are the same clock which provides the sametime, the clocks may have a difference in phase to some extent. In thiscase, it is preferable that the first clock CLK1 lead the second clockCLK2. Also, the third and fourth clocks CLK3 and CLK4 preferably leadthe first and second clocks CLK1 and CLK2.

Note that a delay value of the clock CLK2 to be input to the seconddynamic circuit 1B may not be set to be a predetermined value, and theclock CLK2 may be generated based on the clock CLK3 of the third dynamiccircuit 1C. A circuit structure of this case is illustrated in FIG. 9.In FIG. 9, a dynamic circuit A1 is additionally provided. The dynamiccircuit A1 has a series circuit of the same number of stages of n-typeMOS transistors as the number of stages of n-type MOS transistorsconnected in series in the first dynamic circuit 1A of FIG. 1. Aplurality of the series circuits are connected in parallel to constructa parallel circuit portion, which is the same as that of the firstdynamic circuit 1A, except for a structure of inputting a scan signalSE. An output A1-1 of the dynamic circuit A1 thus additionally providedis inverted in an inverter IN10, and is then input to the n-typetransistor Tr7 of the second dynamic circuit 1B.

The dynamic circuit A1 additionally provided in FIG. 9 further includesa clock generation circuit A2 which generates a clock CLK4 which isinput to the fourth dynamic circuit 1D, based on the clock CLK3 inputfrom the third dynamic circuit 1C of FIG. 1. In the clock generationcircuit A2, a junction capacitance portion of multi-input gates of dataor the like is constructed to be apparently equivalent to the outputpoint A1-1 of the dynamic circuit A1, and an output A2-1 is inverted inan inverter IN11 and is then input to the n-type transistor Tr14 of thefourth dynamic circuit 1D. The clock generation circuit A2 is furtherprovided with a precharge circuit A2-2 comprised of a p-type MOStransistor Tr40. The precharge circuit A2-2 has a function ofprecharging the output point A2-1 of the clock generation circuit A2. Aclock input to the gate of the p-type transistor Tr40 is the clock CLK3which is input to the third dynamic circuit 1C. A delay differenceduring discharge between the output A1-1 of the dynamic circuit A1 andthe output A2-1 of the clock generation circuit A2 depends on a currentdifference between n-type MOS transistors to which the selection signalsS0 to S3 are input. By compensating for the delay difference using theinverter IN11, a reliable operation can be achieved.

Note that, in the circuit of FIG. 1, when any one of the selectionsignals S0 to S3 is output in addition to the selection signal SE, theoutput may become indeterminate if the dynamic circuit A1 istransitioned earlier than the dynamic circuit 1A. However, in FIG. 9,the gates of five NMOS transistors Ts1 to Ts4 which are connected inseries to transistors to which the selection signals SE and S0 to S3 areinput, respectively, are grounded so as not to be conductive. Therefore,since a current path through which the electric charge is dischargedfrom the node A2-1 to the ground is a single path via an NMOS transistorTs5 whose gate is fixed to the power source voltage Vdd, the dynamiccircuit A1 is transitioned later than the dynamic circuit 1A. As aresult, data which is output to the output terminal Q is an OR output ofdata selected from the data D0 to D3. This structure is effective sincean expected value does not become indeterminate when a scan test isperformed.

An exemplary layout structure of the semiconductor integrated circuit ofFIG. 9 is illustrated in FIG. 10. In FIG. 10, a circuit portion ofn-type transistors for receiving the selection signals S0 to S3 of thefirst dynamic circuit 1A and n-type transistors for receiving the dataD0 to D3, and a circuit portion of n-type transistors for receiving theselection signals S0 to S3 of the dynamic circuit A1 of FIG. 9 arevertically arranged. As a result, the wiring capacitance of input pinsis reduced. In addition, since both the circuit portions are close toeach other, a variation component between the dynamic circuits 1A and A1during the production process is reduced, and a voltage variation and atemperature variation are also advantageously reduced. Also, in an inputcircuit portion which is typically comprised of a plurality of n-typetransistors and receives selection signals and data, when a layouthaving a different number of selection signals or pieces of data isproduced, by preparing a layout having a maximum number of inputs, alayout having a smaller number of inputs can be obtained only byreducing the number of n-type MOS transistors on the left side of FIG.10. Therefore, the number of steps for layout can be reduced.

Note that the transistor Tr91 of the dynamic circuit 1A has a functionas a keeper to hold the electric charge of the node N1. In this case, itis desirable that the source of the transistor Tr91 is connected to thedrain (node N20) of the transistor Tr93 of the dynamic circuit A1.Thereby, for example, the signal transition speed of the node N1 becomeshigher than when the source of the transistor Tr91 is connected to thedrain of the transistor Tr94 of the dynamic circuit 1A. This is becausethe drain capacitance of the transistor Tr93 of the dynamic circuit A1does not have an influence on the node N1. The same is true of thetransistor Tr92 of the dynamic circuit 1B.

In addition, when the number of pieces of data to be input isconsiderably large, the pieces of data may be divided into two groups.For example, in a semiconductor integrated circuit of FIG. 11, a groupof the first to fourth dynamic circuits 1A to 1D and A1 of FIG. 9 andanother group of first to fourth dynamic circuits 1A′ to 1D′ and A1′having the same structure as that of the former group are provided. Whenthe number of pieces of data is 2N, data D1 to SN are input to onegroup, while data SDN+1 to D2N are input to the other group. The twogroups are input in parallel to the gates of the n-type transistors Tr20and Tr21 of the output circuit 1E of FIG. 1. Further, a selection signalmatching detection circuit 1J which detects a match between the outputsA1-1 and A1-1′ of the dynamic circuit A1 or a match between the outputsA2-1 and A2-1′ of the clock generation circuit A2 is further provided.An output 1J-1 of the output circuit 1E of FIG. 1 is connected to thegate of the n-type transistor Tr22. With this structure, thecapacitances of the first nodes N1 and N1′ of the first dynamic circuits1A and 1A′ are half of a value which is obtained when only one group isprovided, thereby making it possible to increase the operating speed.

Embodiment 2

Next, Embodiment 2 of the present invention will be described. InEmbodiment 2, the output circuit 1E of FIG. 1 is modified as shown inFIG. 5.

Specifically, an output circuit 1G of FIG. 5 is comprised of adifferential circuit 70. More specifically, the differential circuit 70has a pair of differential input terminals 70 a and 70 b, a pair ofdifferential output terminals 70 c and 70 d, two p-type MOS transistorsTr30 and Tr31 and two n-type MOS transistors Tr32 and Tr33 which arecross-linked, and two n-type MOS transistors Tr34 and Tr35 for receivinga differential signal, to whose gates the pair of differential inputterminals 70 a and 70 b are connected. The differential output terminals70 c and 70 d are connected to a connection point of the p-typetransistor Tr30 and the n-type transistor Tr32 and a connection point ofthe p-type transistor Tr31 and the n-type transistor Tr33, respectively.The differential output terminals 70 d and 70 c are the output terminalQ and its inverted output terminal NQ of the semiconductor integratedcircuit of FIG. 1, respectively.

A signal of the second output node N2 of the second dynamic circuit 1Bof FIG. 1 is input to the differential input terminal 70 a. An ORcircuit 71 is connected to the differential input terminal 70 b. Asignal obtained by inverting the signal of the second output node N2 ofthe second dynamic circuit 1B using an inverter 72, and a signal of thefirst output node N1 of the first dynamic circuit 1A are input to the ORcircuit 71.

Further, a control transistor Tr36 including an n-type MOS transistor isconnected to a ninth node N9 which is the source of the two n-type MOStransistors Tr34 and Tr35 for receiving the differential signal. Thesource of the control transistor Tr36 is grounded, the drain thereof isconnected to the ninth node N9, and the gate thereof receives, as acontrol signal, a fifth clock signal CLK5 which is generated by a clockgeneration circuit 1H of FIG. 6.

An internal structure of the clock generation circuit 1H will bedescribed. In FIG. 6, the clock generation circuit (signal generationcircuit) 1H comprises a short pulse generation circuit 75 whichgenerates a short pulse at the same cycle as that of the first clockCLK1, and a NAND circuit NAND11. The short pulse generation circuit 75has an inverter IN10 which inverts a first clock CLK1, a NAND circuitNAND10 which receives outputs of the first clock CLK1 and the inverterIN10, and an inverter IN11 which inverts an output of the NAND circuitNAND10. The NAND circuit NAND11 receives an output of the inverter IN11and a signal of the fourth output node N4 of the fourth dynamic circuit1D of FIG. 1. An output of the NAND circuit NAND11 is a fifth clockCLK5. The clock CLK5 is input as a control signal to an n-typetransistor Tr36 which is provided in the differential circuit 70 of FIG.5 to receive a differential signal.

In the clock generation circuit 1H of FIG. 6, as illustrated in FIG. 7,for example, it is assumed that the selection signal S1 is High so thatthe data D1 is selected during the first period of the first clock CLK1.Since a signal of the fourth output node N4 of the fourth dynamiccircuit 1D is High at the beginning of the first period, when a shortpulse is subsequently generated by the short pulse generation circuit75, the fifth clock CLK5 then goes from High to Low. Thereafter, whenthe short pulse is ended, the fifth clock CLK5 goes from Low to High. Inthis case, by turning ON the control transistor Tr36 partway throughtransition of the fifth clock CLK5 from Low to High, a differentialinput signal is amplified and output. In the other situations, thecontrol transistor Tr36 is held OFF. Therefore, when the fifth clockCLK5 is High, the output circuit 70 functions as a latch which holdsoutput data. With this structure, when the output circuit 1G of FIG. 5is provided, the holding circuit 1F of FIG. 1 is not required after theoutput circuit 1G.

In FIG. 5, an n-type MOS transistor Tr37 is disposed in parallel withthe control transistor Tr36 in the output circuit 1G. The source of then-type transistor (resistance element) Tr37 is grounded, and the gateand drain thereof are connected to the ninth node N9 of the differentialcircuit 70. There is a possibility that the potential of the ninth nodeN9 is increased due to leakage current when the fifth clock CLK5 is Low.In fact, the n-type transistor Tr37 functions as a resistance element,thereby suppressing and preventing the increase of the potential to holdan optimum potential of the ninth node N9. As a result, the potentialdifference between the source and drain of the n-type transistors Tr34and Tr35 for receiving a differential input is held to be an optimumwhich provides an appropriate gain, whereby a predetermined high-speedoperation of the output circuit 1G is satisfactorily maintained.

In this embodiment, the differential circuit 70 rapidly amplifies andoutputs a small potential difference between input differential signals,thereby operating with higher speed than when data is held by the outputcircuit 1E in Embodiment 1. Note that the differential circuit 70 may bereplaced with a differential amplification circuit 28A of FIG. 28. Inthis case, the gates of P-type transistors 28L and 28M of thedifferential amplification circuit 28A are connected to a clock CLK5.

Embodiment 3

FIG. 8 illustrates a semiconductor integrated circuit according toEmbodiment 3 of the present invention.

The semiconductor integrated circuit of Embodiment 3 is different fromthe semiconductor integrated circuit of FIG. 1 in a first dynamiccircuit 2A of NOR type and a third dynamic circuit 2C of NOR type, andboth the circuits have the same second and fourth dynamic circuits 1Band 1D, output circuit 1E and holding circuit 1F.

In the first dynamic circuit 2A, the p-type transistor Tr1 and then-type transistor Tr2 are connected in series. To this series circuit,an n-type MOS transistor Tr80 which receives data D through the gatethereof, and another n-type MOS transistor Tr81 which receives aninverted signal NQ of the output signal Q through the gate thereof areconnected in series. Therefore, in the first dynamic circuit 2A, thepotential of the first output node N1 is basically determined, dependingon the value of the data D. When the data D is output through the outputterminal Q, the inverted output NQ of the data D is used to handle achange in the next data D.

The third dynamic circuit (matching detection circuit) 2C includes anEXNOR circuit EXNOR1. The EXNOR circuit receives the data D, the outputsignal Q, and inverted signals ND and NQ thereof. After rising of thethird clock CLK3, only when there is a match between the data D and theoutput signal Q, the third output node N3 is set to be the power sourcepotential Vdd. Therefore, in the fourth dynamic circuit 2D, when thereis a match between the data D and the output signal Q, the n-typetransistor Tr13 is turned ON, so that the electric charge of the fourthoutput node N4 is discharged. As a result, the n-type transistor Tr6 isturned OFF in the second dynamic circuit 2B.

With the structure, in the dynamic NAND circuit 2D, when the value ofthe data D is the same as that of the output signal Q, the output nodeN4 goes to Low, so that the n-type transistor Tr6 of the second dynamiccircuit 2B is forcedly turned OFF. Therefore, subsequently, it ispossible to stop operations of the following second dynamic circuit 2B,output circuit 1E and holding circuit 1F. Therefore, unnecessaryoperations of the circuits 2B, 1E and 1F are prevented, thereby makingit possible to reduce the power of the semiconductor integrated circuit.

Note that the physical arrangement of the dynamic circuits, the size andthreshold voltage characteristics of each transistor, voltages suppliedto the circuits, and the like in Embodiment 3 can be similar to those inEmbodiment 1. Further, the output circuit 1E can be replaced with thedifferential output circuit 1G of Embodiment 2. In this case, a stillhigher speed can be achieved.

Although this embodiment illustrates an exemplary flip-flop, a latchcircuit may be implemented by, for example, causing the potential of thenode N2 to be an output signal. In this case, the holding circuit 1Fdoes not have to output a signal or does not have to be provided.

Embodiment 4

FIG. 12 is a circuit diagram illustrating another multi-input flip-flopaccording to the present invention. The multi-input flip-flop of FIG. 12is different from those of FIGS. 1 and 9 in that the flip-flop of FIG.12 is operated with a single clock signal CLK1, and further, in that theflip-flop of FIG. 12 comprises a p-type MOS transistor 12B and a p-typeMOS transistor 12C.

In FIGS. 1 and 9, p-type MOS transistors (transistors Tr4, Tr12 inFIG. 1) are provided whose sources are connected to a power source andwhich are used to charge the nodes N2 and N4. In the circuit of FIG. 12,a p-type MOS transistor 12B is provided whose source and drain areconnected to nodes N1 and N2, respectively, and a p-type MOS transistor12C is provided whose source and drain are connected to nodes N1 and N4,respectively. The gate of the p-type MOS transistor 12B is connected toa node A1-2. The gate of the p-type MOS transistor 12C is connected to anode A2-3. This circuit employs only one clock signal, thereby making itpossible to reduce power consumption, and avoid an erroneous operationdespite use of only one clock signal.

FIGS. 13 and 14 illustrate a relationship between voltage and time ofeach node where, in the circuit of FIG. 12, a signal input patterndiffers between terminals SI, D[1] to D[N−1] and a terminal D[N] orbetween terminals SE, S[1] to S[N−1] and a terminal S[N]. In addition,FIGS. 13 and 14 illustrate waveforms occurring in the circuits of FIGS.1 and 9 when the transistor balance is poor and the circuit is drivenwith a single clock signal, resulting in an erroneous operation. Dashdot lines indicate waveforms when the circuit of FIG. 12 is used, andsolid lines indicate waveforms when the circuits of FIGS. 1 and 9 areused.

A description will be provided in comparison with FIG. 12. In FIG. 13,all input signals of the terminals D[1] to D[N−1], SI, S[1] to S[N], andSE satisfy desired setup and hold times at the timing of transition ofthe clock signal CLK1 to High and are Low. Only the terminal D[N]satisfies desired setup and hold times and is High. Thereafter, during aperiod when the clock signal CLK1 is High, only the terminal S[N] goesfrom Low to High. As a result, nodes A1-1 and N1 go to Low, and a nodeN6 goes to High. When the p-type MOS transistor 12C has a structuresimilar to that of FIGS. 1 and 9, a power source voltage Vdd is suppliedvia the p-type MOS transistor 12C to the node N4 during subsequenttransition of the clock signal CLK1 from High to Low, so that the nodeN4 goes to High. As a result, the High periods of the node N4 and thenode N6 may overlap. When the High periods of the node N4 and the nodeN6 overlap, both transistors Tr21 and Tr22 are caused to be conductive,so that electric charge is discharged from a node N7. In this case,although the node N7 is normally supposed to be held High, the node N7may conversely go to Low, so that an output terminal Q operateserroneously. This is particularly because measures are not particularlytaken in a circuit which controls charge of the node N4 and charge ofthe node N1, so that the node N4 is charged earlier than the node N1,depending on variations in p-type MOS transistor devices which chargethe nodes N4 and N1, leading to an erroneous operation.

In the circuit of FIG. 12, however, current-voltage characteristics of avoltage difference between the drain and source of the p-type MOStransistor 12C exhibit linearity up to near a voltage threshold voltageVtp. Since a substrate voltage of the p-type MOS transistor 12C ishigher than a source voltage thereof, the p-type MOS transistor 12Cbehaves as if it were a considerably high resistance element. Therefore,it is likely that the node N1 is charged before the node N4 is charged.In this case, the timing of transition of the node N4 to High isdelayed, so that the possibility that the nodes N4 and N6 simultaneouslygo to High is reduced.

A further description will be provided in comparison with FIG. 12. InFIG. 14, at the timing of transition of the clock signal CLK1 to High,the terminal S[N] satisfies desired setup and hold times and is High,while input signals of the terminals S[1] to S[N−1], SE, D[1] to D[N],and SI satisfy desired setup and hold times and are Low. Thereafter,during a period when the clock signal CLK1 is High, only the terminalD[N] goes from Low to High. Therefore, the node N1 goes from High toLow. When the p-type MOS transistor 12B has a structure similar to thatof FIGS. 1 and 9, the nodes N1 and N2 are charged during subsequenttransition of the clock signal CLK1 from High to Low. In this case, ifthe node N1 is charged later than the node N2, the node N2 goes to Highwhile the node N1 goes to Low, so that the node N6 goes to High,resulting in a glitch in the node N7. If the glitch is propagated to theoutput terminal Q, an erroneous operation may occur.

In the circuit of FIG. 12, however, current-voltage characteristics of avoltage difference between the drain and source of the p-type MOStransistor 12B exhibit linearity up to near a voltage threshold voltageVtp. Since a substrate voltage of the p-type MOS transistor 12B ishigher than a source voltage thereof, the p-type MOS transistor 12Bbehaves as if it were a considerably high resistor. Therefore, the nodeN2 goes to High only after the node N1 goes to High. Therefore, the nodeN6 does not go to High, thereby preventing an erroneous operation.

As described above, when the source and drain of the p-type MOStransistor 12B are connected to the nodes N1 and N2, respectively, andthe source and drain of the p-type MOS transistor 12C are connected tothe nodes N1 and N4, respectively, the charging order of the nodes N1and N2 and the charging order of the nodes N1 and N4 are uniquelydetermined without depending on manufacturing variations in device sizeof the p-type MOS transistor, thereby making it possible to obtain amore robust circuit structure.

The circuit of FIG. 12 is further characterized in that, in a dynamiccircuit A1, the gates of MOS transistors AN and A3 to AN-1, which areconnected directly to the power source and the ground in FIG. 9, areconnected to two outputs of a circuit 12A.

In the circuit 12A, an n-type MOS transistor 12A-1, a p-type MOStransistor 12A-2, and another n-type MOS transistor 12A-3 are provided.The drain of the p-type MOS transistor (potential setting transistor)12A-2 is connected to the gate of an n-type transistor AN of a secondgroup of n-type transistors A3 to AN to set the gate potential of then-type transistor AN to have a power source potential. Also, the sourceof the n-type MOS transistor 12A-3 is grounded, and the gate and drainthereof are connected to the gate of the potential setting transistor12A-2.

In a miniaturization process, a thickness of a gate oxide film becomesthin, so that the ESD robustness of the gate is reduced. Therefore, inthe circuit of FIG. 9, when an overvoltage is applied to the powersource or the ground, the low impedance is highly likely to causepunchthrough in the gate electrode, likely leading to destruction of theMOS transistor. However, by providing the circuit 12A as illustrated inFIG. 12, the gate of the MOS transistor is connected via a resistancebetween the source and the drain to the power source and the ground.Therefore, there is a high impedance between the gate and the powersource or the ground, whereby the MOS transistor is unlikely to bedestroyed.

It is also preferable that the circuit 12A be provided as a part of themulti-input flip-flop in the same standard cell in which the output ofthe circuit 12A is input to the gates of the second group of n-typetransistors A3 to AN which in turn operate. This is because such amulti-input flip-flop has a number of input terminals, so that wiringbetween standard cells is complicated. Unless the circuit 12A isprovided in the cell, a cell, such as the circuit 12A, needs to beprovided elsewhere, and the cell and the multi-input flip-flop need tobe connected via wiring, so that the degree of Wiring congestion betweenstandard cells is increased. Wiring between standard cells is typicallyautomatically installed. Therefore, wiring may be accidentally arrangedsuch that an output of the circuit 12A is influenced with a crosstalk.When the output of the circuit 12A is contaminated with crosstalk noise,the flip-flop circuit having a multi-input selection function mayperform an erroneous operation. Therefore, in consideration of aninfluence of the crosstalk, it is preferable that the circuit 12A beprovided in the standard cell as long as it is permitted.

In the circuit 12A, the drain of a p-type MOS transistor 12A-2 isassumed to be a node which is connected to the gate of an n-type MOStransistor 12A-1 for the purpose of reduction of the number of devices.Alternatively, similar to the structure of the MOS transistors 12A-2 and12A-3, another p-type MOS transistor is provided, and the drain and gateof the p-type MOS transistor are connected in common to the gate of then-type MOS transistor 12A-1.

When the circuit 12A is provided further below the right and left n-typeMOS transistors in a lower portion of FIG. 10, the circuit 12A can beconnected to the following stage without long wiring of a circuit A1 andthe node N1 of FIG. 12. If the circuit of FIG. 12 is a standard cell,NWELL and PWELL are provided at a lower end thereof again. Therefore,cells can be arranged without considering a distance constraint of aninterface between different wells at an interface between lower cells.

Embodiment 5

FIG. 15 is a circuit diagram illustrating another multi-input flip-flopaccording to the present invention.

The multi-input flip-flop of FIG. 15 is different from those of FIGS. 1and 9 in the flip-flop of FIG. 15 is operated with a single clock signalCLK1, and further, in a circuit portions 13B of the first dynamiccircuit 1E, a circuit portion 13C of the dynamic circuit 1D, and acircuit portion 13A of the dynamic circuit 1A. In FIG. 1, p-type MOStransistors (transistors Tr4, Tr12 in FIG. 1) are provided whose sourcesare connected to a power source and which are used to charge the dynamicnode portions N2 and N4. In the circuit of FIG. 15, further, otherp-type MOS transistors (p-type MOS transistor 13B1, p-type MOStransistor 13C1) are connected to the drain of a p-type MOS transistorfor charging, and are connected via the source and drain to nodes N2 andN4, respectively. The gate of the p-type MOS transistor 13B1 and thegate of the p-type MOS transistor 13C1 are connected to an output of aninverter circuit INV13 of the node N1. Further, although the source ofthe p-type MOS transistor 13A is connected to the power source in FIG.11, it is connected to a node A1-1 of FIG. 15. Thus, this circuitemploys only one clock signal, thereby making it possible to reducepower consumption and avoid an erroneous operation despite use of onlyone clock signal.

Also, in FIG. 15, an n-type transistor (first n-type transistor) Tr40 isprovided in the third dynamic circuit A1. A clock signal CLK1 is inputto the gate of the n-type transistor Tr40, and the sources of the n-typetransistors (the second group of n-type transistors) A3 to AN areconnected in common to the drain thereof. Further, the common source ofa plurality of n-type transistors (third group of n-type transistors)A20 to AK is connected to the common drain of the second group of n-typetransistors A3 to AN. On the second group of n-type transistors A3 toAN, a predetermined power source is connected to the gate of the n-typetransistor AN so that the gate potential is set to be a power sourcepotential. Also, the gates of the other n-type transistors A3 to A5 areall grounded so that the gate potentials are set to be a groundpotential. The selection signals S[1] to S[N] are input to the gates ofthe third group of n-type transistors A20 to AK, respectively, and athird output node N3 is connected in common to the drains of the thirdgroup of n-type transistors A20 to AK.

An inverted node A1-2 of the third output node N3 (=A1-1) of the thirddynamic circuit A1 is connected to the second dynamic circuit 1E. Aninverted node A2-2 of the node A2-1 of the common drain of the secondgroup of n-type transistors A3 to AN as well as the third output node N3are connected to the fourth dynamic circuit 1D.

FIGS. 16 and 17 illustrate a relationship between voltage and time ofeach node where, in the circuit of FIG. 15, a signal input patterndiffers between terminals D[1] to D[N−1] and a terminal D[N] and betweenterminals S[1] to S[N−1] and a terminal S[N]. In addition, FIGS. 16 and17 illustrate waveforms occurring in the circuits of FIG. 9 when thetransistor balance is poor and the circuit is driven with a single clocksignal, resulting in an erroneous operation. Dash dot lines indicatewaveforms when the circuit of FIG. 15 is used, and solid lines indicatewaveforms when the circuit of FIG. 9 is used.

A description will be provided in comparison with FIG. 15. In FIG. 16,all input signals of the terminals S[1] to S[N] satisfy desired setupand hold times at the timing of transition of the clock signal CLK1 toHigh and are Low. Thereafter, during a period when the clock signal CLK1is High, only the terminal S[N] goes from Low to High. As a result,nodes A1-1 and N1 go to Low, and a node N6 goes to High. When thecircuit 13C has a structure similar to that of FIGS. 1 and 9, a powersource voltage Vdd is supplied via two p-type MOS transistors 13C1 and13C2 to the node N4 during subsequent transition of the clock signalCLK1 from High to Low, so that the node N4 goes to High. As a result,the High periods of the node N4 and the node N6 may overlap. When theHigh periods of the node N4 and the node N6 overlap, both transistorsTr21 and Tr22 are made conductive, so that electric charge is dischargedfrom a node N7. In this case, although the node N7 is normally supposedto be held High, the node N7 may conversely go to Low, so that an outputterminal Q operates erroneously. This is because measures are notparticularly taken in a circuit which controls charge of the node N4 andcharge of the node N1, so that the node N4 is charged earlier than thenode N1, depending on variations in p-type MOS transistor devices whichcharge the nodes N4 and N1, leading to an erroneous operation.

In the circuit of FIG. 15, however, the circuit 13C is not turned ONunless a potential of an output of the inverter circuit INV13 of thenode N1 is smaller than or equal to a difference between a thresholdvoltage of a p-type MOS transistor in the circuit 13C and a power sourcevoltage Vdd. Therefore, it is likely that the node N1 is charged earlierand the node N4 is charged later. Therefore, the possibility that thenodes N4 and N6 are simultaneously High is reduced.

A further description will be provided in comparison with FIG. 15. InFIG. 17, at the timing of transition of the clock signal CLK1 to High,the terminal S[N] satisfies desired setup and hold times and is High,while input signals of the terminals S[1] to S[N−1], SE, D[1] to D[N],and SI satisfy desired setup and hold times and are Low. Thereafter,during a period when the clock signal CLK1 is High, only the terminalD[N] goes from Low to High. Therefore, the node N1 goes from High toLow. Thereafter, in the circuit of FIG. 1, the nodes N1 and N2 arecharged during subsequent transition of the clock signal CLK1 from Highto Low. In this case, if the node N1 is charged later than the node N2,the node N2 goes to High while the node N1 goes to Low, so that the nodeN6 goes to High, resulting in a glitch in the node N7. If the glitch ispropagated to the output terminal Q, an erroneous operation may occur.

In the circuit of FIG. 15, however, the node N2 is not charged unless apotential of an output of the inverter circuit INV13 of the node N1 issmaller than or equal to a difference between a threshold voltage of ap-type MOS transistor 13B1 in the circuit 13B and the power sourcevoltage Vdd. Therefore, the node N2 goes to High only after the node N1goes to High. Therefore, the node N6 does not go to High, therebypreventing an erroneous operation.

Further, in FIG. 18, when the clock signal CLK1 goes to High, theterminals D[N] and S[N] satisfy desired setup and hold times and areHigh, while input signals of the terminals S[1] to S[N−1], SE, D[1] toD[N−1], and SI satisfy desired setup and hold times and are Low.Thereafter, during a period when the clock signal CLK1 is High, theterminal D[N] goes from High to Low. Thereafter, the clock signal CLK1goes from High to Low. In this case, the node A1-1 and the node N1 arecharged. The node N1 reaches a threshold voltage Vtn of the n-type MOStransistor earlier than the node A1-1, depending on transistorvariations in the p-type MOS transistor. In this case, a through currentflows through the node N25, resulting in a glitch in the node N2. Theglitch is propagated to the node N7, so that an erroneous operationoccurs in the output terminal Q.

In the circuit of FIG. 15, however, since the source of the p-type MOStransistor 13A is connected to the node A1-1, current-voltagecharacteristics of a voltage difference between the drain and source ofthe p-type MOS transistor 13A exhibit linearity up to near a voltagethreshold voltage Vtp. Since a substrate voltage of the p-type MOStransistor 13A is higher than a source voltage thereof, the p-type MOStransistor 13A behaves as if it were a considerably high resistor.Therefore, the node A1-1 is charged first before start of charging ofthe node N1. Therefore, after a gate voltage of an n-type MOS transistor1E-1 becomes smaller than or equal to a threshold voltage of the n-typeMOS transistor, a gate voltage of an n-type MOS transistor 1E-2 becomeseasier to be larger than or equal to a threshold voltage, so that thethrough current of the node N2 becomes difficult to flow, and therefore,a glitch does not occur in the node N7. Further, in FIG. 15, the gate ofa p-type MOS transistor 13B2 and the gate of a p-type MOS transistor13C2 are connected to the clock signal CLK1.

In the circuit of FIG. 12, discharging of the node N2 starts only aftera voltage of a node A1-2 reaches (Vdd-Vtp) or more. In FIG. 15, when theclock signal CLK1 has a voltage of (Vdd-Vtp) or more, the node N2 isready for discharging. Therefore, advantageously, the node N2 canoperate faster than that of FIG. 12.

As described above, the source of the p-type MOS transistor 13B2 isconnected to the power source. The drain of the p-type MOS transistor13B2 is connected to the source of the first p-type MOS transistor 13B1.The drain of the p-type MOS transistor 13B1 is connected to the node N2.The gate of the second p-type MOS transistor 13B2 is connected to theclock signal CLK1. The gate of the p-type MOS transistor 13B1 isconnected to the output of the inverter circuit INV13 of the node N1.The source of the p-type MOS transistor 13C1 is connected to the powersource. The drain of the p-type MOS transistor 13C1 is connected to thesource of the p-type MOS transistor 13C1. The drain of the p-type MOStransistor 13C2 is connected to the node N4. The gate of the p-type MOStransistor 13C2 is connected to the clock signal CLK1. The gate of thep-type MOS transistor 13C1 is connected to the output of the invertercircuit INV13 of the node N1. The source of the p-type MOS transistor13A is connected to the node A1-1. As a result, the charging order ofthe node A1-1 and the node N1, the charging order of the node N1 and thenode N2, and the charging order of the node N1 and the node N4 are eachuniquely determined without depending on manufacturing variations indevice size of the p-type MOS transistor, thereby making it possible toobtain a more robust circuit structure.

The structure in which the source of the p-type MOS transistor 13A isconnected to the node A1-1 is described above. Alternatively, the sourceof the p-type MOS transistor 13A may be connected to the drain ofanother p-type MOS transistor, whose source may be in turn connected tothe power source and whose gate may be in turn connected to the outputof the inverter circuit of the node A1-1. In this case, a similar effectcan be obtained. In other words, the present invention may beimplemented with any circuit structure in which the charging order ofthe node A1-1 and the node N1, the charging order of the node N1 and thenode N2, and the charging order of the node N1 and the node N4 can eachbe uniquely determined without depending on manufacturing variations indevice size of the p-type MOS transistor. Such a circuit structure canbe achieved with a combination of various circuits, and does not departfrom the scope of the present invention.

Embodiment 6

FIG. 19 is another circuit diagram illustrating the dynamic circuits 1Cand 1D of the multi-input flip-flop of FIG. 1.

The dynamic circuits 1C and 1D of FIG. 19 are different from those ofFIG. 1 in that first and second p-type MOS transistors A13 and N14A areprovided in place of the p-type MOS transistor Tr9 for charging the nodeN3; the clock signal CLK3 is input to the gate of the p-type MOStransistor N14A, the source of the p-type MOS transistor N14A isconnected to a power source, and the drain of the p-type MOS transistorN14A is connected to the node A2-2 (i.e., the common source of the thirdn-type transistors A20 to AK); and the source and drain of the otherp-type MOS transistor A13 are connected to the nodes N3 and the nodeA2-2 (i.e., the common drain and the common source of the third n-typetransistors A20 to AK), respectively. Further, although the clock signalCLK4 is connected to the gate terminal of the transistor Tr14 of thedynamic circuit 1D in FIG. 1, an output of an inverter circuit IN14 isconnected to the gate terminal of the transistor Tr14 in FIG. 19.

Such a circuit structure has an advantage such that, when a signalhaving the same phase as that of the clock signal CLK3 is input to theclock signal CLK4, i.e., the circuit is driven only based on the clocksignal CLK3 as in FIG. 19, the circuit can be operated with an evenlower power source voltage. The reason will be described as follows. Inthe circuit structure of FIG. 1, it is assumed that the clock signalCLK4 and the clock signal CLK3 having the same phase are input. When theclock signal CLK3 goes from Low to High with a low power source voltagewhich is in the vicinity of a threshold voltage of the n-type MOStransistor (e.g., the threshold voltage of the n-type MOS transistor is0.3 V and the power source voltage is 0.5 V), it takes an overwhelminglylonger time for the node N3 to discharge than for the gate terminal ofthe transistor Tr14. In this case, the node N4 goes to Low, but notHigh, though the transistor Tr13 is normally supposed to be cut off andthe node N4 is normally supposed to go to High (i.e., any of theterminals S[1] to S[N] and the terminal SE goes to High).

In the structure of FIG. 19, however, when the clock signal CLK3 goesfrom Low to High, the nodes N3 and A2-2 simultaneously startdischarging. When the node N14A goes to no more than a switching levelof the inverter circuit IN14, the voltage of the gate of the transistorTr14 is increased. Therefore, the node N3 goes to no more than thethreshold voltage of the n-type MOS transistor Tr13 before the gate ofthe transistor Tr14 goes to High. In this case, it is unlikely that athrough current flows through the node N4 via the transistors Tr13 andTr14. As a result, a low-voltage operation is stabler than that of thecircuit structure of FIG. 1.

Further, when the clock signal CLK3 goes from High to Low,current-voltage characteristics of a voltage difference between thedrain and source of the p-type MOS transistor A13 exhibit linearity upto near a voltage threshold voltage Vtp. Since a substrate voltage ofthe p-type MOS transistor A13 is higher than a source voltage thereof,the p-type MOS transistor A13 behaves as if it were a considerably highresistor. The node N3 is charged only after the potential of the nodeA2-2 becomes higher than or equal to the threshold voltage of the p-typeMOS transistor A13. In other word, the transistor Tr13 is turned ON onlyafter the gate of the transistor Tr14 is lowered to some extent. Sincethe node N4 is charged in accordance with the clock signal CLK3, aglitch is suppressed from occurring in the potential of the node N4 whenthe transistor Tr13 is turned ON. As a result, an erroneous operationwhich is involved with the dynamic the circuits A1 and 1D is suppressed.

Embodiment 7

FIG. 20 illustrates an exemplary application of FIG. 11.

In FIG. 11, a flip-flop is provided which has a multi-input selectionfunction in which input data is divided into two groups. In FIG. 20,transistors of the output circuits 1E are combined to construct a NANDlogic circuit with respect to outputs of a multi-input selectionfunction comprised of dynamic circuits 1A to 1D, and A1 and amulti-input selection function comprised of dynamic circuits 1A′ to 1D′,and A1′.

Specifically, two p-type MOS transistors Tr20 are provided which have acommon source and drain. Two p-type MOS transistors Tr21 are connectedin series to each other. Further, in a holding circuit 50 which iscomprised of a first inverter INV15 connected to the drain of the p-typeMOS transistor Tr20 and a second inverter INV16 which receives an outputof the inverter INV15, one stage of n-type MOS transistor 16A to thegate of which the second output nodes N2 of the dynamic circuits 1A to1D and A1 are connected is provided between a P-type MOS transistor Tr60and a N-type MOS transistor Tr61 constituting the second inverter INV16(this structure is similar to that of FIG. 11). In FIG. 20, one stage ofn-type MOS transistor 16B to the gate of which the second output nodesN2′ of the dynamic circuits 1A′ to 1D′ and A1′ are connected isprovided, thereby maintaining the high speed of the holding circuit 50.Note that the two stages of n-type MOS transistors may be providedbetween the ground and the n-type MOS transistor Tr61 constituting thesecond inverter INV16.

In this embodiment, an exemplary NAND logic has been described. Thepresent invention is not limited to this. Various combined logiccircuits can be produced. In addition, by replacing a dynamic logicportion involved in the dynamic circuit 1A or 1A′ with various logics, aflip-flop circuit having more various combined logic functions can beconstructed. Further, by connecting an N-type transistor between Tr5 andthe node N2 of FIG. 1 and connecting the output of the dynamic circuit1A having another logic to the gate of the N-type transistor, morevarious functional logics can be constructed. Furthermore, a MOStransistor may be added to the transistor Tr20 or the transistor Tr21,and the gate terminal thereof may be connected to an output of anothermulti-input dynamic circuit. The resultant circuit does not depart fromthe scope of the present invention.

FIG. 21 illustrates another exemplary application of FIG. 11, in whichthe source and drain of a transistor Tr21 in each output circuit areconnected in common.

Embodiment 8

FIG. 22 illustrates another exemplary application of FIG. 11, in whichonly a scan input circuit is provided in dynamic circuits 1A′ to 1D′ andA1′.

The dynamic circuits 1A′ to 1D′, A1′, 17B and 17C are static flip-flopswhich share a holding circuit portion 17E and an output portion of anoutput terminal Q with a multi-input selection function flip-flopcomprised of dynamic circuits 1A to 1D and A1. Further, the circuit ofFIG. 22 is different from that of FIG. 11 in that the gate of an n-typeMOS transistor 17D is connected to an inverted output of a scan enablesignal SE. The static flip-flop may be circuits as illustrated in FIGS.23 and 24.

With this circuit structure, when the scan enable signal is activated,the transistors Tr22 and Tr20 are cut off, while only the circuitelements 17B and 17C are operated. The circuit has an advantage suchthat the capacitance of the node N1 can be reduced, and in an ordinarypath, high speed can be achieved by using a dynamic flip-flop. For ascan path, a hold time during scan input is shortened by using a staticflip-flop, thereby effectively securing a margin for a scan shiftoperation.

Note that, by combining an output circuit portion of the dynamic circuitand an output portion of the static circuit with an output circuitportion 17F, a flip-flop circuit having more various logic functions canbe obtained. As described above, in the present invention, theadvantages of the dynamic circuit and the static circuit can beselectively utilized, depending on the function of an input signal or adesired specification.

The eight embodiments have been heretofore described. It is easy forthose skilled in the art to exchange a portion of the circuit structureof a semiconductor integrated circuit of any one of the eightembodiments with a portion of the circuit structure of any one of theother seven embodiments. For example, the dynamic circuit 1B of FIG. 8may be exchanged with the dynamic circuit 1B of FIG. 9.

Hereinafter, specific examples in which the above-describedsemiconductor integrated circuit is applied to predetermined circuitswill be described in Embodiments 9 to 11 below.

Embodiment 9

FIG. 25 illustrates an embodiment in which the flip-flop of Embodiment 1or Embodiment 8 is applied to a data path of a processing unit as apredetermined circuit.

FIG. 25 illustrates a data path 25A, a memory 25J, and a register file25K. The data path 25A has a three-stage pipeline structure. The firststage includes 10 multi-input flip-flops 25Z of the embodiment above,each of which has 14 data input per bit of data. Outputs of themulti-input flip-flops 25Z are input to ALUs 25B1 to 25B3, a bypass unit25C1, a convolution 25E1, a divider 25F1, and a multiplier 25G1. Thesecond stages includes forwardings 25D1 to 25D3, a bypass unit 25C2, aconvolution 25E2, a divider 25F2, and a multiplier 25G2. The thirdstages includes forwardings 25D4 to 25D6, and a multiplier 25G3.Normally, data output from the register file 25K is selected by theflip-flop 25Z. When a data hazard occurs in a pipeline process, aforwarding path is used to avoid the pipeline process from beingdisturbed. The number of pieces of data per bit of the forwarding pathis 13 which is the sum of the number of pieces of output data from thepipeline stages and the number of data output lines 25L from a memory.Data which has been subjected to arithmetic processing in the ALU or thelike needs to be input to the flip-flop. Therefore, in order to achievehigh-speed pipeline processing, a time required to set up data inputfrom the forwarding path of the flip-flop is caused to be as small aspossible. In the flip-flop described in the embodiment above, the datasignal setup time is substantially zero (when the delay of an inverteris, for example, 45 psec where the fan-out is 4, a setup time of, forexample, 300 psec is required in a flip-flop with a static logicselector control circuit illustrated in the conventional example. In theflip-flop described in the embodiment above, the data setup time is, forexample, 10 psec, and the control signal setup time is, for example, 30psec), and therefore, a higher speed is achieved than that ofconventional static logic structures, resulting in higher pipelineprocessing speed.

Also, even when the data path pipeline is stalled, the flip-flop of thepresent invention is useful. A logic for controlling a data pathrequires a control circuit which, when a pipeline in the data path isstalled, compares addresses of data and determines which data is to beselected in the next cycle, so as to determine whether or not loadeddata can be effectively used. However, in the flip-flop of the presentinvention, the control signal setup time is also substantially zero, andtherefore, a higher speed is achieved than that of conventional staticlogic structures, resulting in higher pipeline processing speed.Further, a data control can be performed using a transistor size smallerthan when a selection signal is arranged by conventional static logics,thereby making it possible to arrange data paths with a small area. Inaddition, in static logics, a data transition delay time of each inputvaries due to a wire resistance or the like, so that a glitch occurs inan output of each static logic, and useless power is generated untildata is established. In the configuration of the flip-flop of thepresent invention, each data and its control line are directly connectedto a data input port and a control input port of the flip-flop withoutvia a static logic path, whereby useless power is not generated.

Embodiment 10

FIG. 26 illustrates an embodiment in which the flip-flop described inthe embodiment above is used in a crossbar bus switch, which is used ina system-on-chip or the like.

In FIGS. 26, 26A indicates a processor core and 26B indicates a DMA. 26Cis an SDRAM interface block which interfaces with an SDRAM outside thechip. 26D indicates a system bus interface block which interfaces with aROM, a memory or the like outside the chip. 26E indicates an on-chipmemory interface block which controls interface with an on-chip memoryor a co-processor. 26F indicates an on-chip I/O interface block whichcontrols interface with an on-chip I/O. The crossbar bus switch hasthree masters (not shown), two of which are provided in the processorcore 26A and one of which is provided in the DMA 26B. The crossbar busswitch also has four slaves (not shown), which are the SDRAM interfaceblock 26C, the system bus interface block 26D, the on-chip memoryinterface block 26E, and the on-chip I/O interface block 26F. Betweenfour slave buses 26G and each master, a four-input data selectioncontrol circuit 26J and a flip-flop 26I which receives its output signalare present. As the four-input data selection control circuit 26J, themulti-input flip-flop of the present invention is used. Thereby, sincesetup times for data and a control signal are short in the multi-inputflip-flop, the transfer rate of the bus can be improved by a reductionin the setup time. Further, since the value of the flip-flop can be heldwhen no control signal is selected, the control of data bus arbitrationis not required in each master, resulting in a structure with a smallarea.

Note that, by using the flip-flop of the present invention with respectto the input to the slave, an effect similar to the above-describedmaster input can be obtained.

Embodiment 11

FIG. 27 illustrates an embodiment in which the flip-flop of the presentinvention is applied to a reconfigurable processor.

In FIGS. 27, 27A indicates each processor element in the reconfigurableprocessor. The processor element 27A includes a multi-input flip-flop27C comprised of the flip-flop described in the embodiment above, anoperator 27D, a register file 27E, and the like. 28A indicates buses inthe reconfigurable processor. Data output from one processor element 27Ais directly connected to four data buses. Also, data inputs of themulti-input flip-flop 27C in the processor element 27A are connected tothe four respective data buses. In the reconfigurable processor, some ofthe processor elements 27A are simultaneously subjected to an operationprocess so as to improve the processing performance of a certainapplication. For example, when a processor element group 27F isseparated from other processor element groups 27G, 27H and 27I,processor elements in each group perform data communication with eachother. In this case, since the four groups are separated from eachother, an independent bus interface can be constructed in each group. Byusing the multi-input flip-flop of the present invention, a staticselection control logic as in the conventional art is not required,resulting in a high-speed bus interface in each group and a small areaof each processor element. Further, regarding clocks of the elementgroups 27G, 27H and 27I, when the elements 27A in the element group areconnected in parallel to perform parallel processing, a clock having thesame phase is used, and when the elements 27A in the element group areconnected in series to perform serial processing, clocks havingdifferent phases are used. For example, when the element group 27Iperforms two-stage serial processing, a clock having a phase differenceof 0 may be used for the first-stage element 27A, while a clock having aphase difference of 180 degrees may be used for the second-stageelement. Thereby, by stopping the clock in the second-stage element 27Aand providing a static data selection circuit which bypasses theflip-flop, the effect of reducing the area and increasing the speed isobtained. Also, with the above-described structure, even when a certainapplication is switched to another different application, so that thegroup structure of processor elements needs to be changed, aconventional static selection control logic is not used, so that acontrol of changing any bus lines can be performed with high speed.Therefore, switching latency is small, resulting in a higher-performancereconfigurable processor.

Embodiment 12

FIG. 28 illustrates a semiconductor integrated circuit according toEmbodiment 12 of the present invention.

This embodiment is different from the semiconductor integrated circuitof FIG. 3( a) in that a differential amplification circuit 28A isinserted between output nodes N1 and N2 of first and second dynamiccircuits 28A1A and 28A1B and a holding circuit 90.

An activation signal 28A0 for the differential amplification circuit 28Ais a signal obtained by slightly delaying the clock CLK of FIG. 3( a)from a resistance element or a capacitance element comprised of asemiconductor device. In this embodiment, the activation signal 28A0 isconnected via an output signal node 28A11 of a dynamic circuit 28A1 andan inverter 28A01 to the gate of the active N-type transistor 28A1 ofthe differential amplification circuit 28A.

The differential amplification circuit 28A comprises the N-typetransistor 28A1, a differential pair transistor in which the gates ofN-type transistors 28B and 28C whose sources are connected to the drainof the N-type transistor 28A1 are input terminals thereof, and a latchcircuit 28D which amplifies a difference in drain voltage between theN-type transistors 28B and 28C. The latch circuit 28D comprises fivetransistors, in which an inverter 28G comprising an N-type transistor28E and a P-type transistor 28F is cross-connected with an inverter 28Jcomprising an N-type transistor 28H and a P-type transistor 281. Also,the drains of the N-type transistors 28B and 28C are connected to eachother via a high-resistance element (here, an NMOS transistor 28K). Theoutput of the inverter 28G is a node OUT, and the output of the inverter28J is a node OUTB. Also, when the activation signal is inactivated, thenodes OUT and OUTB are charged to substantially a power source voltagevalue by the P-type transistors 28L and 28M, respectively.

The node N1 is connected to the gate of the N-type transistor 28B, andthe node N2 is connected to the gate of the N-type transistor 28C.

The holding circuit 90 comprises a P-type transistor 28N having a gateto which the node OUT is connected, and an N-type transistor 28Q havinga gate to which the node OUTB is connected via an inverter 28P, in whichthe drain of the P-type transistor 28N is connected with the drain ofthe N-type transistor 28Q, and the connection node NQ is connected viaan inverter 28R to an output pin Q.

With the structure above, when a voltage difference between the voltageof the output node N1 and the voltage of the output node N2 is small,the voltage difference is amplified by the differential amplificationcircuit 28A with higher speed, so that the voltage difference can bequickly increased to a switching level of the holding circuit. As thenumber of pieces of input data D0 to DN and the number of theirselection signals S0 to SN are increased, the high-speed operationeffect is further exhibited than when the dynamic circuit 28A1 and theholding circuit 90 are directly connected to each other as illustratedin FIG. 3( a). This is because the load capacitance of the node N1 isincreased with an increase in those numbers, and the voltage transitiontimes of the output node N1 and the output node N2 when connected to atypical holding circuit (e.g., the holding circuit of FIG. 1) is delayedsince the voltage transition time of the output signal of the holdingcircuit is proportional to the voltage transition times of the nodes N1and N2.

Note that, in this embodiment, a structure is employed in which theactivation signal 28A0 is transferred via the dynamic circuit 28A1.Therefore, the capacitance of the node of the dynamic circuit 28A1 isfurther reflected when the clock CLK is delayed by several stages ofbuffers. Therefore, an optimal delay value which allows an appropriatevoltage difference between the output node N1 and the output node N2 isachieved and the number of transistors is reduced, so that a smallerarea and lower power consumption are also achieved.

Note that, when the number of input data signals of the dynamic circuit28A1 is small, the clock CLK may be directly used as an activationsignal for the differential amplification circuit 28A.

Also, although, in this embodiment, the dynamic circuit 28A1 of FIG. 3(a) having a function of selecting any of a plurality of data signals isused, a similar effect is obtained even when a dynamic circuit havinganother logic function, such as a dynamic circuit having a function ofpropagating only one data signal or the like, is used.

Further, as illustrated in FIG. 29, the holding circuit 90 may receiveonly one output of the differential amplification circuit 28A and may beconnected to the gates of the N-type transistor 28Q and the P-typetransistor 28N. An N-type transistor 28Q1 to the gate of which theactivation signal 28A0 of the differential amplification circuit 28A isinput may be connected in series between the N-type transistor 28Q andthe ground.

In addition, although, in this embodiment, the differentialamplification circuit 28A is provided in a circuit without the thirddynamic circuit (non-selected state detection circuit) 1C of FIG. 1, thedifferential amplification circuit 28A may be provided in a circuithaving the third dynamic circuit (non-selected state detection circuit)1C of FIG. 1 and the like.

Embodiment 13

FIG. 30 illustrates a semiconductor integrated circuit according toEmbodiment 13 of the present invention. This embodiment is differentfrom the semiconductor integrated circuit of FIG. 28 in that a functionof holding the content of data of the holding circuit 90 when none ofthe selection signals S0 to SN selects the data signals D0 to DN isadded.

A dynamic circuit 28A1C comprises a third dynamic circuit 28A1CA and afourth dynamic circuit 28A1CB. Only selection signals S0 to SN are inputto the third dynamic circuit 28A1CA. An output node A1C-2 of the thirddynamic circuit 28A1CA is input to the fourth dynamic circuit 28A1CB,and an output node of the fourth dynamic circuit 28A1CB is a node N4.When one or more of the selection signals S0 to SN are selected, thenode N4 holds High after a clock goes from Low to High. Also, the nodeA1C-2 goes from High to Low. When none of the selection signals S0 to SNis selected, the node N4 goes from High to Low after the clock goes fromLow to High, and the node A1C-2 holds the High level. When the selectionsignal S0 goes to High after exceeding a desired hold value, the nodeA1C-2 goes from High to Low, while the node N4 remains Low. The nodeAC1-2 and the node N4 are transferred to the gates of two N-typetransistors 28A1 and 28AA connected in series in the differentialamplification circuit 28A. Therefore, when no selection signal isactivated, the differential amplification circuit 28A is not activated,so that the nodes OUT and OUTB remain High, and therefore, the node NQof the holding circuit 90 is not changed.

As described above, a function of holding the content of data of theholding circuit 90 when no selection signal is activated can beachieved. Thereby, when none of a plurality of pieces of data isselected, the holding circuit 90 can hold the previous data informationeven after the clock is transitioned, thereby making it possible toreduce setup times of data and a selection signal as compared to theconventional art.

Also, regarding the dynamic circuit 28A1 and the dynamic circuit 28AC1,since the dynamic circuit 28AC1 has a smaller load capacitance, thedynamic circuit 28AC1 has quicker voltage transition of the output node.Therefore, the differential amplification circuit 28A starts anoperation during transition of the output nodes N1 and N2 of the dynamiccircuit 28A1, so that, even when a voltage difference between the nodesN1 and N2 is small, the voltage difference is amplified with higherspeed than in the amplification circuit 28A. In addition, it isadvantageous that the vertically symmetrical layout as in FIG. 10 doesnot need to be formed.

Although, in this embodiment, the differential amplification circuit 28Ais provided in a circuit without the third dynamic circuit (non-selectedstate detection circuit) 1C of FIG. 1, the differential amplificationcircuit 28A may be provided in a circuit having the third dynamiccircuit (non-selected state detection circuit) 1C of FIG. 1 and thelike.

Embodiment 14

FIG. 31 illustrates a semiconductor integrated circuit according toEmbodiment 14 of the present invention. This embodiment includes theform of the semiconductor integrated circuit of FIG. 11, and further,has a form useful in a case where the vertically symmetrical layout asin FIG. 10 cannot be achieved due to the specification of the height ofa cell when the semiconductor integrated circuit is physically designed(layout design).

In FIG. 31, a dynamic circuit 28A1 has a first dynamic circuit 28A1A anda second dynamic circuit 28A1B, receives data signals D0 to DN andselection signals S0 to SN, and outputs nodes N2 and N1. On the otherhand, a dynamic circuit 28A1C has a third dynamic circuit (non-selectedstate detection circuit) 28A1CA and a fourth dynamic circuit 28A1CB,receives only selection signals S0 to SN, and outputs nodes N4 andA1C-2. In the first dynamic circuit 28A1A, the number of stages ofN-channel transistors connected in series for selecting any of aplurality of pieces of data (i.e., transistors for inputting data D,selection signals S, and a clock CLK) is three. In the third dynamiccircuit (non-selected state detection circuit) 28A1CA, the number ofstages transistors connected in series for detecting a state in whichnone of a plurality of selection signals is selected (i.e., transistorsfor inputting data D, selection signals S, and a clock CLK) is two,which is smaller by one than the number (three) of stages connected inseries in the first dynamic circuit 28A1A.

In FIG. 31, inverted data of the output node N2 and the output node N1are input to a NOR circuit 30D whose node OUT30A is connected to thegate of an N-type transistor 28Q of a holding circuit 90, and the outputnode N2 is connected to the gate of a P-type transistor 28N. Also,inverted data of the output node N4 and the output node A1C-2 are inputto a NOR circuit 30C whose node OUT30B is connected to the gate of anN-type transistor 28Q1, and whose inverted signal OUT30C is connected tothe gate of a P-type transistor 28N1. The N-type transistors 28Q1 and28Q are connected in series, the P-type transistors 28N and 28N1 arealso connected in series, and the drain of the N-type transistor 28Q andthe drain of the P-type transistor 28N are connected in common, and theresultant node is NQ.

An operation of the above-described structure will be described. Whenthe clock CLK is Low, the node N2 of the dynamic circuit 28A1 and thenode N4 of the dynamic circuit 28A1 are High, and the nodes OUT30A andOUT30B are Low.

Data signals D0 to DN and selection signals S0 to SN are input to thedynamic circuit 28A1. When one of the selection signals S0 to SN isselected and the data signals D0 to DN are High, the node N2 holds Highafter the clock goes to Low to High. Also, the node N1 goes from High toLow. OUT30A goes from Low to High.

When none of the selection signals S0 to SN is selected, the node N2goes from High to Low after the clock goes from Low to High, and thenode N1 holds High. The node OUT30A holds Low. After the data signal andthe selection signal exceed desired hold values, when any of theselection signals S0 to SN goes to High, the node N1 goes from High toLow, while the node N2 remains Low. Therefore, the node OUT30A remainsHigh.

Only selection signals S0 to SN are input to the dynamic circuit 28A1C.When one or more of the selection signals S0 to SN are selected, thenode N4 holds High after the clock goes from Low to High. Also, the nodeA1C-2 goes from High to Low. The node OUT30B goes from Low to High.

When none of the selection signals S0 to SN is selected, the node N4goes from High to Low after the clock goes from Low to High since thenode A1C-2 holds High. The node OUT30B holds Low. After the selectionsignal S0 exceeds a desired hold value (e.g., the selection signal S0goes to High), the node A1C-2 goes from High to Low, while the node N4remains Low. Therefore, the node OUT30B remains Low. Therefore, whennone of the selection signals S0 to SN is selected, the holding circuit90 holds data, and when any of the selection signals S0 to SN isactivated, selected data of the data signals D0 to DN is output as anoutput Q.

Regarding the dynamic circuit 28A1 and the dynamic circuit 28A1C, thedynamic circuit 28A1C has a smaller load capacitance, the dynamiccircuit 28A1C has quicker voltage transition of the output node thanthat of the dynamic circuit 28A1. Therefore, before transition of theoutput nodes N2 and OUT30A of the dynamic circuit 28A1, the nodes OUT30Band OUT30C hold transition or a potential. Therefore, when none of theselection signals S0 to SN is activated, data in the holding circuit 90can be reliably held.

As described above, in the dynamic circuit 28A1C, it is not necessary toprovide the vertically symmetrical physical design (layout design) as inFIG. 10 in which a dummy data input transistor is required (the dynamiccircuit 28AC1 does not require a dummy data input transistor), so thatthe number of transistors is also reduced, and therefore, a small areaand low power consumption are possible. Also, in the dynamic circuit28A1C, the charge times of the output node N4 and the node A1C-2 areshorter than that of the output nodes N1 and N2 of the dynamic circuit28A1, and therefore, even when the clock CLK goes from High to Low, aglitch is unlikely to occur in the node NQ of the holding circuit 90, sothat the semiconductor integrated circuit is unlikely to malfunction.

Note that a signal input to the gate of the P-type transistor 28N1 maybe an output of a logic structure which is High when the dynamic circuit28A1C is not selected and is Low when the dynamic circuit 28A1C isselected, and can be generated by various logic structures using thenode N4 and the node A1C-2.

Embodiment 15

FIG. 32 illustrates a semiconductor integrated circuit according toEmbodiment 15 of the present invention. This embodiment is differentfrom that of FIG. 3( a) in that a setup absorption circuit 31A isinserted between the node N1 of the first dynamic circuit 28A1A and thenode N2 of the second dynamic circuit 28A1B and the holding circuit 90.

The setup absorption circuit 31A comprises a switch circuit 31B, acircuit 31C for charging a node N21 and holding a High potential level,and an N-type transistor 31D. The gate of the N-type transistor 31D isconnected to a node N1, the source of the N-type transistor 31D isconnected to a node N2, and the drain of the N-type transistor 31D isconnected to a node N21. Also, the switch circuit 31B comprises atransfer gate. When the node N2 goes to Low, the potential of the nodeN21 is transferred to the node N2 via a buffer 31E comprised of twoinverters 31E1 and 31E2 and the switch circuit 31B. The node N21 ischarged when the node N2 is High.

An operation of the setup absorption circuit 31A will be described withreference to a timing chart of FIG. 33. The horizontal axis representstimes, and the vertical axis represents voltage values of each signal,data D, a clock CLK, the node N1, the node N2, the node N21, and a nodeN22.

In FIG. 33( a), when the clock CLK is Low, the data D is Low, and afterthe clock CLK goes from Low to High, the data D goes from Low to Highslightly later than a prescribed setup time. This case will behereinafter discussed. Since the output node N1 does not comply with thedata setup time, the output node N1 slowly goes from High to Low.Therefore, the output node N2 goes from High to Low. When data arriveswithin the prescribed setup time, the output node N2 is normallysupposed to hold High. Thereafter, the node N22 is inverted. In thetiming chart, while the node N22 is inverted, the output node N1 goes toLow, so that the node N21 is held High. Therefore, the output node N2goes from Low to High via the switch circuit 31B, and the output Q ofthe holding circuit 90 outputs High. In FIG. 33( a), dashed linesindicate when the data D is significantly delayed form the prescribedsetup. In this case, the output node N2 remains Low, and the output Q ofthe holding circuit 90 outputs Low, so that erroneous data is output.

In FIG. 33( b), when the clock CLK is Low, the data D is High, and afterthe clock CLK goes from Low to High, the data D goes from High to Lowslightly later than a prescribed setup time. This case will behereinafter discussed. Since the node N1 does not comply with the setuptime of the data D, the output node N1 slowly goes from High to Low.Normally, the node N1 needs to hold High. Therefore, the node N2 slowlygoes from High to Low. Normally, the node N2 needs to quickly go to Low.In the timing chart, while the output node N1 slowly goes from High toLow, the data D goes to Low, so that a keeper circuit for the outputnode N1 causes the node N1 to gradually go from an intermediatepotential to High. Thereafter, the node N22 is inverted. While the nodeN22 is inverted, the output node N1 goes to High, so that the node N21goes from High to Low. Therefore, the output node N2 remains Low via theswitch circuit 31B, and the output of the holding circuit 90 outputsLow. In FIG. 33( b), dashed lines indicate when the data D issignificantly delayed form the prescribed setup. In this case, theoutput node N2 remains High, and the output Q of the holding circuit 90outputs High, so that erroneous data is output.

As described above, even when the data D slightly violates the setup,the setup absorption circuit 31A allows the output Q to indicate anormal value, resulting in a circuit robust against a variation inprocess, a variation in power source voltage, and the like.

Although, as illustrated in FIG. 34, the source of the N-type transistor31D of the setup absorption circuit 31A is connected to the output nodeN2, the source of the N-type transistor 31D of the setup absorptioncircuit 31A may be connected to the drain of an N-type transistor 33Ewhose source is grounded. Also, a signal line on which the clock CLK isfurther delayed by inverters 33C and 33B may be connected to the gate ofthe N-type transistor 33E. In this case, the dynamic node N21 isconnected to the output node N2 only via a switch element which iscontrolled by the signal line on which the clock is delayed. Thereby,when the data D goes from Low to High slightly later than the prescribedsetup, the node N2 is quickly charged, resulting in high-speedtransition of the voltage of the output Q. Alternatively, the invertedoutput of the output node N2 may be output to the gate of the N-typetransistor 33E.

Also, for the gate of the N-type transistor 33E, a clock which satisfiesa holding time constraint of data with higher precision, and a clockhaving a different phase may be used. Thereby, the violation of thesetup time of the data D can be absorbed until the very end of theholding time of the data D. Such a clock having a different phase may begenerated by distorting the duty ratio (a ratio of a High period to aLow period) of the clock CLK and inverting the resultant clock.

Although, in this embodiment, the dynamic circuit 28A1 having a functionof selecting any of a plurality of data signals using a plurality ofselection signals is used (FIG. 3( a)), a similar effect is exhibited byusing a dynamic circuit having another logic function, such as a dynamiccircuit having a function of propagating only a single data signal orthe like.

Also, although, in this embodiment, the setup absorption circuit 31A isprovided in a circuit without the third dynamic circuit (non-selectedstate detection circuit) 1C of FIG. 1, the setup absorption circuit 31Amay be provided in a circuit having the third dynamic circuit(non-selected state detection circuit) 1C of FIG. 1 and the like.

INDUSTRIAL APPLICABILITY

As described above, in the present invention, even when no selectionsignal is activated, so that no data is selected, the output signal ofthe holding circuit can be satisfactorily held at the previous value.Therefore, the present invention is useful as a dynamic flip-flopcircuit with a data selection function or the like.

Also, in the present invention, when input data already matches thevalue of an output signal from the holding circuit, the operation of atleast a portion of the dynamic flip-flop circuit can be forcedlyinterrupted. Therefore, the present invention is preferably applied to asemiconductor integrated circuit in which a useless operation issuppressed to achieve lower power consumption, or the like.

1. A semiconductor integrated circuit, comprising: a dynamic logiccircuit having a plurality of first input lines for data signals and afirst output line; a scan input circuit having a second input line for adata signal and a second output line; and a holding circuit having afirst transistor, a second transistor, and a third output line, whereinthe first output line is coupled to a gate of the first transistor ofthe holding circuit, and the second output line is coupled to a gate ofthe second transistor of the holding circuit.
 2. A semiconductorintegrated circuit, comprising: a dynamic logic circuit having aplurality of first input terminals for data signals and a first outputterminal; a scan input circuit having a second input terminal for a datasignal and a second output terminal; and a holding circuit having afirst transistor, a second transistor and a third output line, wherein afirst output from the first output terminal of the dynamic logic circuitis input to a gate of the first transistor of the holding circuit, and asecond output from the second out put terminal of the scan input circuitis input to a gate of the second transistor of the holding circuit.
 3. Asemiconductor integrated circuit, wherein the semiconductor integratedcircuit of claim 1 is a flip-flop circuit.
 4. A semiconductor integratedcircuit, wherein the semiconductor integrated circuit of claim 2 is aflip-flop circuit.